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Abstract
An electrically tunable pattern-reconfigurable integrated array antenna based on a 1-bit digital coding metasurface is proposed in this paper. The array antenna consists of 8 × 8 unit cells which is divided into the radiation antenna and the phase control metasurface. The rectangular microstrip patch antenna is used as the radiation source. The phase control metasurface includes three stacked layers of square slot units loaded with varactor diodes and a biasing circuit layer. The metasurface regulates the transmission phase of the quasi-plane wave generated by the radiation antenna. The pattern reconfigurability is achieved by switching coding sequences of the metasurface which can be regulated by the capacitance of varactor diodes. The measured results show the design has the capability of pencil-beam radiation, beam deflection, and multi-beam radiation at 5 GHz. The proposed array antenna preserves the coding metasurface without requiring the additional horn antenna. This highly integrated design is more concise and reduces the profile height. This article offers what we believe is a new method for the design of high integration, multi-unit, and electrically tunable reconfigurable array antenna.
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1. Introduction
With the rapid development of wireless communication systems and the application of multiple input multiple output (MIMO) communications, compact reconfigurable antennas with high integration and high gain are attracting more and more attention. In recent years, reconfigurable antennas with the capability of switching polarization [1], frequency [2], and radiation pattern [3,4] have played an important role in 5 G wireless communication systems. Especially, pattern reconfigurable antenna is capable of dynamically switching radiation patterns, which has the advantages of improving system performance and decreasing the interference of noise. In the designs [5–7], the activation states of radiation patches can be respectively excited by the on-off state of PIN diode loaded into the feeding network to generate numerous radiation patterns, such as single-directive and multi-beam pattern, switchable broadside and conical beams pattern. The beam pattern can be directionally scanned in the horizontal plane. However, due to the complexity of the feeding network and the size of the antenna element, these designs consist of just one or several elements, which limits the gain and pattern reconfigurable capability.
As the special two-dimensional form of metamaterials, metasurface can regulate the amplitude and phase of electromagnetic wave flexibly [8], and also has the advantage of low profile, simple design, and low insertion loss. In 2014, the concept of coding metasurface was proposed [9,10]. For 1-bit digital coding metamaterial, two types of unit cells with a phase difference of 180° are encoded as “0” and “1”. In this digital way, the specific encoding sequence of the digital “0” and “1” elements can ensure the digital coding metasurface has various regulatory functions on electromagnetic wave [11–13]. These interesting functionalities include beam steering [14,15], radar cross-section reduction [16], imaging [17], and wireless power transfer [18]. However, the introduction of the additional primary feeding source (mostly horn antenna) will increase the profile height of the system, reduce integration and affect the aperture efficiency of the system. Therefore, the antenna that integrates the feeding source with the metasurface is proposed which realizes the arbitrary control of the phase and amplitude of the radiation wave [19]. One design integrates the Fabry–Perot antenna with the metasurface [20], and another design directly excites parts of the low-RCS metasurface cells to produce efficient radiations [21]. The integrated antenna can realize reconfigurable polarization [22], holographic imaging [23], reconfigurable frequency [24], and reconfigurable radiation pattern [25,26], while having the advantages of simple feeding network, low profile, high integration, and high gain.
In this paper, we propose a high gain and pattern reconfigurable integrated array antenna based on binary coding metasurface. The array antenna operates at 5 GHz consists of 8 × 8 unit cells, and it is divided into the radiation antenna and the phase control metasurface. The rectangular microstrip patch antenna is used as the radiation antenna, and the phase control metasurface is three stacked layers of square slot unit loaded with varactor diodes. The metasurface regulates the transmission phase of the quasi-plane wave generated by the radiation antenna to realize pencil-beam radiation, beam deflection, and multi-beam radiation, by electrically adjusting the capacitance of varactor diodes.
This paper is organized as follows. Section 2 presents the detailed structure and operating mode of the radiation antenna and the phase control metasurface. Section 3 presents the theoretical calculation and simulation result of the far-field radiation pattern. In Section 4, a prototype array antenna is fabricated and the experimental results are compared with the simulation results. Finally, conclusions are drawn in Section 5.
2. Integrated antenna unit design
2.1 Radiation antenna
The unit of the integrated array antenna designed in this paper is multi-layer structure, which is divided into two parts: the phase control metasurface (PCM) and the radiation antenna (RA), as illustrated in Fig. 1. To reduce the profile height of the integrated antenna, the rectangular microstrip patch antenna is used as the radiation source. The vertically polarized wave is along the z-direction. And the square slot PCM loaded with varactor diodes is designed to regulate the transmission phase of the electromagnetic wave excited by the RA.
Fig. 1. The integrated antenna element consists of the phase control metasurface and the radiation antenna.
Figure 2(a) shows the dimensions of the patch radiation antenna and Fig. 2(b) shows the side view of the radiation antenna. It is composed of three layers: the patch layer, the ground plane, and the feeding network layer. The 1 mm diameter metal via is used to connect the top patch layer to the bottom feeding network layer without connecting to the ground. The geometries of the radiation antenna unit are chosen as p = 33 mm, w = 25.55 mm, L = 15.8 mm, H1 = 2 mm, H2 = 1.524 mm, and the thickness of copper is 0.035 mm. The patch layer substrate is chosen as F4BM (${\varepsilon _r}$ = 2.5 and tan $\delta $ = 0.002) and the feeding layer substrate is Rogers RO4350B (${\varepsilon _r}$ = 3.66 and tan $\delta $ = 0.0037). The characteristic impedance of the via is 127 Ω, which is determined by the diameter. The reflection coefficient of each unit varies with the position of the via. To reduce reflection, the S-parameter is calculated when the port impedance is 127 Ω to obtain the optimized position of the via, and it finally determines the distance y0 between the via and the center is 4.6 mm.
Fig. 2. (a)The detailed structure of the radiation antenna unit. (b) The side view of the radiation antenna unit.
To increase the gain, we choose an 8 × 8 radiation patch antenna array, the multi-layer structure is shown in Fig. 3(a). The impedance of the array feed port is determined as 50 Ω. All patch units are connected to the 50 Ω feed port through a matching circuit based on the microstrip transmission line, the detailed structure is shown in Fig. 3(b). The dimensions are chosen as w1 = 1.4 mm, w2 = 2.3 mm, w3 = 5.17 mm, w4 = 8.59 mm, and L1 = 33 mm. The symmetrical feeding network has 64 output ports and each port corresponds to a patch through the via. The distance between the output ports is equal to the size p of the patch antenna. The feeding network uses graded impedance conversion to feed all units with the same amplitude and phase. The perforated ground plane isolates the patch and the feeding network, reducing the influence of the microstrip transmission line on the radiation beam.
Fig. 3. (a) The multi-layer of the radiation antenna array and (b) the detailed structure of the feeding layer.
The radiation antenna array is simulated and optimized in the CST Studio Suite 2020 which was developed by the Germany Computer Simulation Technology AG. The simulated S-parameter and radiation pattern are shown in Fig. 4(a) and Fig. 4(b). We can confirm that the center resonant frequency of the antenna unit is 5 GHz with a working bandwidth of 80 MHz. Its E-plane radiation pattern is symmetrical, and the main beam is a standard pen beam with the maximum gain of 21 dBi at 5 GHz.
2.2 Phase-control metasurface
The phase control metasurface composed of 8 × 8 equal-sized unit cells is designed to be integrated with the radiating antenna array to regulate the radiation phase of each unit independently. The phase control metasurface includes three stacked layers of square slot units loaded with varactor diodes and an additional feeding network layer. The geometry parameters of the PCM unit cell are as follows: p = 33 mm, d = 25 mm, w = 3 mm, g = 1 mm, as depicted in Fig. 5(a). The PCM unit is also loaded on the substrate F4BM with a thickness of 1.5 mm.
Fig. 5. (a)The detailed structure of the phase control metasurface unit. (b) The equivalent circuit of the phase control metasurface unit.
The varactor diodes are loaded in the slot, the positive electrode and negative electrode are connected with the square ring and the patch respectively. The center of the patch is fed by the fourth feeding layer through a 1 mm diameter via so that the diode capacitance can be controlled by its reverse voltage. The unit cell is also simulated and optimized in the CST Studio Suite. The unit cell exhibits a band-pass like frequency response with phase control to vertically polarized incident electromagnetic waves. This corresponding resonant behavior can be analyzed by the equivalent LC circuit, as illustrated in Fig. 5(b), and the resonant frequency is calculated as follows,
The simulated transmission amplitude and phase of the single-layer PCM unit are shown in Fig. 6(a) and Fig. 6(b) respectively. We can observe that, as the capacitance decreases from 2.8 pF to 0.8 pF, the band-pass characteristics of the unit cell shift from 4.08 GHz to 4.58 GHz and the phase difference increases gradually. The transmission amplitude of a single-layer PCM unit can reach 0.7 or above within a working frequency range of 4.33 GHz to 4.43 GHz with a bandwidth of 100 MHz, and the transmission phase difference exceeds 60° when the diode operates in two states with the capacitors of 2.8 pf and 0.8 pf, respectively.
For 1-bit digital coding metasurface units, it is necessary to obtain the maximum transmission amplitude on the basis of satisfying the 180° transmission phase distribution. Generally, we use the digital code “0” and “1” to denote two different elements whose transmission phase is $\psi $ and $\psi $ + π, and $\psi $ is an arbitrary phase. We designed different code sequences to control the transmission phase shifting of PCM, so as to achieve different beam-forming effects. Three layers of unit cells are stacked and separated by a 2 mm air gap to extend the range of the transmission phase. To reduce the profile height as much as possible, the thickness of the substrate and the air gap have been compressed to a minimum. The simulated amplitude and phase of the multi-layer PCM unit are shown in Fig. 7(a) and Fig. 7(b). The transmission phase difference of the three layers element is 150° at 4.48 GHz, which conforms to the requirement of the phase transmission range of 180° ± 30°. Although there exists a slight frequency shifting compared with the single-layer unit, the radiation amplitude of the PCM unit can exceed 0.7 in two working states at the operating frequency, meaning that the PCM unit has a relatively high transmission efficiency. We choose the unit with the diode capacitance of 2.8 pF as digital code “0” and the unit with the diode capacitance of 0.8 pF as digital code “1”. The conversion between code “0” and “1” is achieved by loading corresponding voltages.
We design coding sequences of the PCM composed of 8 × 8 units to generate different radiation beams. The square rings of all units are connected as the negative electrode, which means the voltage loaded on all units uses the same negative electrode. The positive electrode via in the center of each unit connects the three stacked layers and finally connects to the pad of the biasing circuit. This feeding method allows each unit can be independently regulated by the bias voltage. The multilayer structure is shown in Fig. 8(a).
Fig. 8. Detailed structure of (a) the phase-control metasurface (layer 1-4) and (b) the biasing circuit (layer 4).
Since the metasurface is 1-bit coded, we select four typical coding sequences and design the corresponding biasing network. The biasing circuit is symmetrical and uses narrow microstrip lines to transmit direct current. The pad with label is the feeding port, and the pad in the middle corresponds to 64 PCM units, as shown in Fig. 8(b). We use eight ports to control all the 64 units. Each port can feed eight elements simultaneously. For example, port 1 controls units 1, 2, 5, 6 in row 1 and row 2. Port 2 controls units 1, 2, 5, 6 in row 3 and row 4. Port 3, 4 have similar functions, they control units 1, 2, 5, 6 in the remaining rows. Port 5, 6, 7, 8 are the mirror of port 1, 2, 3, 4, they control units 3, 4, 7, 8 in different rows respectively. The digital code of the unit is “0” when the capacitance of the diode is 2.8 pF, which is achieved by loading 0.88 v DC voltage. And digital code “1” is achieved by 8.33 v voltage.
To get the coding sequences S1 (00 00 00 00) or S2 (11 11 11 11), we feed the DC voltage 0.88 v or 8.33 v to all ports. The coding sequence S3 (00 11 00 11) can be obtained by feeding 0.88 v voltage to ports 1-4 and 8.33 v voltage to ports 5-8. And the coding sequence S4 (00 11 00 11 /11 00 11 00) can be obtained by feeding 0.88 v and 8.33 v to ports 1, 3, 6, 8 and ports 2, 4, 5, 7 respectively. Therefore, we can realize arbitrary switching of the coding sequences by feeding different DC voltages to the corresponding ports of the bias circuit. Compared to individual PCM units, the PCM array exhibits better radiation performance at 5 GHz rather than at 4.48 GHz. Therefore, the operating frequency of the antenna array is determined to be 5 GHz.
3. Radiation patterns simulation
The electrically tunable beam steering characteristic of the integrated antenna is obtained by integrating the radiating antenna with the PCM, the multi-layer structure is shown in Fig. 9. The high gain radiation antenna array provides a quasi-planar wave to the PCM, which is responsible for controlling the phase of the transmission wave to obtain the designed far-field radiation pattern. Therefore, the radiation performance of the integrated antenna is affected by the air gap between the radiation source and the PCM. To reduce the profile height of the integrated antenna, the air gap needs to be decreased as much as possible, while a narrow air gap will produce coupling which will affect the radiation beam. We confirm that the integrated antenna has high radiation performance when the air gap is decreased to 10 mm by parameter sweeping.
As we know, the far-field radiation pattern is calculated by the coding sequence of “0” and “1” elements. Take an N × N binary metasurface array with uniform element spacing p for instance, the shaped field pattern $f({\theta ,\phi } )$ of the transmitted field under plane wave excitation can be expressed as [4],
Just as the calculation, the integrated array antenna with the coding sequence S1 (00 00 00 00) or S2 (11 11 11 11) radiates a single beam along the z-axis. The polarization of the beam is linear with a gain of 16.7 dBi, as illustrated in Fig. 10(a).
Fig. 10. Simulated far-field radiation pattern of the integrated array antenna at 5 GHz. (a) coding sequence S1 (00 00 00 00). (b) coding sequence S3 (00 11 00 11). (c) coding sequence S4 (00 11 00 11/11 00 11 00).
By changing the bias voltage, the coding sequence is converted to S3 (00 11 00 11) and the far-field radiation of the integrated antenna is converted into dual-beam radiation. In such a case, λ = 60 mm at 5 GHz, ${\Gamma _x}$ = 4 × p = 132 mm, and ${\Gamma _y}$ → ∞ because the coding sequence has no change along the y-direction. According to formulas (2) and (3), the elevation angle θ is calculated as 27° and the azimuth angle $\phi $ is 0°. We can observe from Fig. 10(b) that the dual-beam far-field radiation simulation result with each lobe gain of 12.5 dBi is located in the x-z plane and the elevation angle is 25°. The simulation result is in good agreement with the calculation. For the coding sequence “S4 (00 11 00 11/11 00 11 00)”, the integrated antenna radiates four centrosymmetric main beams in the upper-half space. Substituting ${\Gamma _x}$ = ${\Gamma _y}$ = 132 mm into (2) and (3), the theoretical elevation angle θ is calculated as 40° and the azimuth angles $\phi $ are ± 45° and ± 135°. As shown in Fig. 10(c), the elevation angle θ and the azimuth angle $\phi $ of the radiation beams are (37.5°,45°), (36.9°,135°), (37.6°,225°) and (36.7°,315°), respectively. The gain of each beam is 9.7 dBi which has a decrease of 2.8 dBi in contrast to the coding sequence S2. And the S-parameters of the integrated antenna with different coding sequences are shown in Fig. 11, we can observe that the reflection coefficient of the integrated antenna has a slight frequency offset after integration with the PCM, but the integrated antenna still has a great radiation performance at 5 GHz.
Fig. 11. Simulated S-parameter of the integrated array antenna with coding sequences S1, S3, and S4.
4. Experimental verification
An integrated beam-forming array antenna sample was fabricated to validate the design. Four 8 mm diameter holes were made at the corners of all floors to integrate the layers with bolts and isolation gaskets. The size of the sample is 300 mm × 300 mm × 26 mm. Figure 12(a) shows the top view of the PCM and Fig. 12(b) illustrates the bottom view of the radiation antenna feeding network. To reduce the cost, the varactor diode SMV1232-079LF from SKYWORKS is selected for capacitance changing range and low price. According to the datasheet, the intrinsic parameters of the varactor diode such as the diode’s series inductance (0.7 nH) and series resistance (1.5Ω) are considered in the simulation to compensate for frequency deviation that may occur in the actual situation. The capability of electrically tunable beam-forming on the integrated antenna is confirmed by testing the S-parameter and far-field radiation properties in the microwave anechoic chamber. The testing environment is shown in Fig. 12(c).
Figure 13 shows the simulated and measured S-parameter and gain of the radiation antenna array without the PCM. We can observe that the measured reflection coefficient of the radiation antenna array operates at 5 GHz, and the gain of the radiation antenna can reach 19 dBi. The simulation S-parameter operates at 5.025 GHz, and the frequency difference between simulation and measurement is 0.025 GHz, which can be negligible. The efficiency of the radiation antenna array can reach 73% at 5 GHz. The results demonstrate that the radiation antenna array has good radiation performance within the operating frequency range.
Fig. 13. Measured and simulated (a) S-parameter and (b) the gain of the radiation antenna array without the PCM.
Figure 14 shows the measured S-parameters of the integrated array antenna with designed coding sequences. It is observed that compared with the radiation antenna array without PCM, the S11 of the integrated array with designed coding sequences has a slight frequency deviation, but the integrated antenna still has a good matching effect at 5 GHz.
Figure 15 shows the normalized far-field radiation pattern of the integrated array antenna with the coding sequences S1/S2, S3, and S4 at 5 GHz. Figure 15(a) shows the integrated antenna with the coding sequence S1 radiates a standard pencil beam with a gain of 12.38 dBi. Due to the actual dielectric loss of the substrate which is larger than the simulation, as well as measurement error and fabrication tolerances, the measured gain is reduced compared with the simulated gain. The radiation pattern of the integrated array is transformed into dual-beam located in the x-z plane with a gain of 9.2 dBi when the coding sequence is converted from S1 to S3, and the measured elevation angle of the radiation beam is 24°, which is very close to the simulation result 25°, as shown in Fig. 15(b). Figure 15(c)-(d) shows the far-field pattern of the integrated array with coding sequence S4, dual-beam with the gain of 7.5 dBi can be seen in two planes with the azimuth angles 45° and 135° respectively. The measured elevation angle $\theta $ of the main beam is 36° which has a good agreement with simulation.
Fig. 15. The normalized far-field radiation pattern of the integrated array antenna at 5 GHz. (a) radiation pattern of the integrated array antenna with the coding sequence S1 (00 00 00 00) and (b) x-z plane radiation pattern of the coding sequence S3 (00 11 00 11). (c) phi = 45° and (d) phi = 135° plane radiation patterns of the integrated array antenna with the coding sequence S4 (00 11 00 11/11 00 11 00).
Compared to other integrated antenna [19], this design realizes the real-time reconfiguration of the coding sequences and the radiation patterns. And compared to the traditional metasurface or Fabry-Perot antenna [23], this design reduces the profile height and improves the integration without horn or resonator cavity. In the future research, this design method can extend the coding sequences to 2-bit coding to obtain a wider beam scanning range.
5. Conclusion
In conclusion, an electrically-tunable pattern-reconfigurable integrated array antenna composed of radiation antenna and 1-bit coding metasurface is presented. The proposed integrated antenna consists of 8 × 8 unit cells loaded with varactor diodes, and the coding sequences can be regulated instantly by the electrically controlled capacitance of varactor diodes. A prototype of the integrated antenna whose coding sequence is electrically controlled by the feeding network is fabricated, and the measured radiation pattern can be switched between single-beam, dual-beam, and multi-beam instantly at 5 GHz which has a good agreement with simulation. The proposed integrated antenna preserves the coding metasurface for real-time control of electromagnetic wave and increases integration without requiring traditional horn antenna as the feeding source. This may offer a new method for the design of high integration, multi-unit, and pattern-reconfigurable array antenna.
Funding
National Key Research and Development Program of China (2020YFA0709800); National Natural Science Foundation of China (61971340, 62122061); Shaanxi Natural Science Basic Research Project of Shaanxi science and Technology Office (2023-JC-QN-0673).
Disclosures
The authors declare that there are no conflicts of interest related to this article.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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