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Abstract
We report a photonic approach to generate background-free multi-format dual-band microwave signals based on a single modulator, which is suitable for high-precision and fast detection of radars in complex electromagnetic environments. By applying different radio-frequency signals and electrical coding signals to the polarization-division multiplexing Mach-Zehnder modulator (PDM-MZM), the generation of dual-band dual-chirp signals or dual-band phase-coded pulse signals centered at 10 and 15.5 GHz is experimentally demonstrated. Furthermore, by choosing an appropriate fiber length, we verified that the generated dual-band dual-chirp signals are not affected by chromatic-dispersion induced-power fading (CDIP); meanwhile, by autocorrelation calculations, we got high pulse compression ratios (PCRs) of 13 of the generated dual-band phase-encoded signals, showing that the generated phase-encoded signals can be emitted directly without extra pulse truncation operation. The proposed system features a compact structure, reconfigurability and polarization independence, which is promising for multi-functional dual-band radar systems.
© 2023 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
Pulse compression is widely used in modern radars to increase the range resolution [1]. Thanks to their excellent pulse compression ratio, linearly chirped signals and phase-coded signals have been the most common radar waveforms for large-range, high-resolution target detection [2,3]. Traditionally, linearly chirped signals and phase-coded signals are generated by electrical methods such as the analog mixer and direct digital synthesizer. However, limited by the electronic bottleneck, the generated linearly chirped signals and phase-coded signals cannot meet the demands of modern radars with larger bandwidth and higher carrier frequency. Therefore, the photonic generation of linearly chirped signals and phase-coded signals has been investigated in recent years, taking full advantages of microwave photonics technology such as large bandwidth, high carrier frequency, low loss, electromagnetic interference (EMI) immunity and so on [4–9].
Unfortunately, linearly single-chirped signals have a knife-edge-type ambiguity function, which will degrade the range-Doppler resolution and detection accuracy. To overcome the problem, dual-chirp signals as a promising candidate are proposed for radar systems, which consist of complementarily linearly chirped signal pairs. Various photonic schemes have been reported to generate dual-chirp signals [10–17]. In [10,11], dual-chirp signals with frequency and bandwidth multiplying were generated based on a dual-polarization dual-parallel Mach-Zehnder modulator (DP-DPMZM) or two cascaded MZMs. Considering that the dual-chirp signal generated in the central office (CO) needs to be transmitted to remote base stations (BSs) in multi-static radar, the chromatic-dispersion induced-power fading (CDIP) suppression is a crucial technique. Hence, other photonic methods have been presented to generate dual-chirp signals with fiber dispersion elimination [12–14]. In addition, the multi-band radar system has attracted lots of attention recently due to its versatility such as multispectral imaging, large coverage weather monitoring, immunity to complex EMI and so on. Therefore, the photonic generation of multi-band dual-chirp signals has been researched in [15–17] for multi-band radar applications.
Many photonic schemes have been put forward to generate phase-coded signals [18–24]. In [19–21], single-frequency phase-coded pulse signals were generated successfully based on a dual-polarization MZM (DPol-MZM) or a DP-DPMZM. Similarly, to be suitable for multi-band radars, multi-band phase-coded signals have been achieved in [22–24] by balanced detection, polarization selection or nonlinearity of electro-optic modulation. However, in most existing works, the generated multi-band phase-coded signals are in continuous wave (CM) mode, which need to be truncated into pulses by intensity modulation before being emitted by radar antenna, resulting in background noise. Recently, we have reported a photonic scheme to directly obtain multi-band phase-coded pulses by cascaded DP-DPMZM and polarization modulator (PolM), which is complicated. Therefore, the photonic generation of multi-band phase-coded pulse is charming in multi-band radar applications.
To increase the multi-scenario applicability of radar systems, photonic systems that can generate multi-format switchable microwave signals have been investigated [25–27]. In [25], a photonic-assisted microwave signal generator was proposed to generate linearly chirped signals and phase-coded signals. To further improve the detection accuracy of radars, a photonic approach was reported to generate multi-band switchable dual-chirp signals and phase-coded signals based on a DPol-MZM, which is affected by background noise and sensitive to environmental changes due to balanced photodetection [26]. Hence, another photonic method was presented to generate background-free dual-chirp signals and phase-coded pulse signals, which is single-frequency and not suitable for multi-band radars [27]. Therefore, it is desirable to generate multi-band switchable background-free dual-chirp signals and phase-coded pulse signals.
In this paper, we propose a compact photonic system to generate multi-format dual-band microwave signals based on a single polarization-division multiplexing Mach-Zehnder modulator (PDM-MZM). By properly setting the format of the applied baseband signal and the direct current (DC) biases of the PDM-MZM, background-free dual-band dual-chirp microwave signals or phase-coded pulse signals can be generated successfully, which averts unwanted baseband crosstalk. Moreover, the generated dual-band dual-chirp signals are free from the chromatic-dispersion induced-power fading (CDIP), which is suitable for one-to-multi-base station fiber transmission; the generated dual-band phase-coded signals are in pulse mode, which can be emitted directly by radar antennas without extra pulse truncation operation. The proposed scheme distinguishes itself from those previously described in the literature [13,14] by its ability to generate dual-band dual-chirp signals, while previous schemes were limited to generating such signals within a single band. Furthermore, we have demonstrated that our scheme can also generate dual-band phase-encoded pulse signals, which has not been reported in [17]. In addition, the proposed system is simple and flexible, which has great potential for multi-functional and multi-scenario radar and communication systems.
2. Principle
2.1 Fundamentals of microwave photonics links
The schematic diagram of the proposed multi-format dual-band microwave signals generator is shown in Fig. 1, which consists of a laser diode (LD), a PDM-MZM, a tunable optical band-pass filter (TOBPF), an erbium-doped fiber amplifier (EDFA), a spool of single-mode fiber (SMF) and a photodetector (PD). A continuous linearly polarized optical light from LD is injected into a PDM-MZM, which is mainly composed of two dual-drive MZMs (MZM1 and MZM2) at two orthogonal polarization states, a 90-degree polarization rotator (90° PR) and a polarization beam combiner (PBC). On the one hand, the baseband signal Vs·s(t) from the arbitrary waveform generator (AWG) is divided into two paths by a power divider, and then sent to one radio-frequency (RF) port of MZM1 and MZM2, respectively. On the other hand, two different RF signals V1·cos(ω1 t) and V2·cos(ω2 t) from the microwave source (MS) are respectively connected to the other RF ports of the two MZMs. By setting the DC bias voltages of both MZMs to work at the maximum transmission point, the optical fields output by MZM1 and MZM2 can be expressed as
Fig. 1. Schematic diagram of the proposed photonic-assisted multi-format dual-band microwave signal generator without background noise. LD: laser diode, EA: electric amplifier, MZM: Mach-Zehnder modulator, AWG: arbitrary waveform generator, 90° PR: 90° polarization rotator, PBC: polarization beam combiner, TOBPF: tunable optical band-pass filter, EDFA: erbium-doped fiber amplifier, SMF: single-mode fiber, PD: photodetector, OSA: optical spectrum analyzer, ESA: electrical spectrum analyzer, DPO: digital phosphor oscilloscope.
To remove the optical sidebands higher than the first-order sideband, we cascade a tunable optical band-pass filter (TOBPF) behind the PDM-MZM. Meanwhile, by adjusting the amplitudes of RF1 and RF2 are adjusted to make β1=β2 = 2.4, that is J0(β1)=J0(β2) = 0, as shown in Fig. 2. Therefore, Eq. (2) can be simplified as
After the power compensation of the output signal of the PDM-MZM by the EDFA, two optical signals with orthogonal polarization states are injected into the PD for photoelectric conversion. The recovered photocurrent is given by
As can be seen from the Eq. (4), aside from the frequency doubled microwave components and the desired chirp components, the DC components of the generated signal are constant. This implies that there is no baseband-modulated signals present in the DC components, and hence, the experimental setup is free from background noise.
2.2 Generation of dual-band dual-chirp signals for anti-dispersion transmission
When the baseband signal is a parabolic microwave signal, viz. s(t)=k(t-T0/2)2, where T0 is the duration of the s(t) signal, and k is the chirp rate, which is 4/T02. Ignoring the DC and high-frequency components, Eq. (4) can be re-expressed as
It can be seen from Eq. (5) that the proposed scheme can successfully generate dual-band dual-chirp microwave signals without the influence of background noise. When the signal is transmitted from the CO to the BS through the single-mode fiber, as shown in Fig. 1, the optical field at the output port of the SMF can be expressed as
According to Eq. (8), the dispersion-induced phase shifts only affect the phase of the generated dual-chirp signal instead of its amplitude. Therefore, the dual-chirp signal generated based on this scheme can realize anti-dispersion transmission for radar networks with one-to-many BSs. Note that the bandwidth of the generated dual-chirp signal is 4Vs/VπT0, and the time-bandwidth product (TBWP) is 4Vs/Vπ. Considering that Vπ is fixed, by splitting the parabolic microwave signal into N pieces with equal amplitude, the TBWP of the generated signal can be further increased to 2NVs/Vπ [13].
2.3 Generation of dual-band phase-coded pulse signals
When the baseband signal s(t) is a phase-encoded signal, ignoring the DC and high-frequency components, Eq. (4) can be re-written as
According to Eq. (9), this scheme can realize the generation of a dual-band phase encoding signal, and the polarity of the encoded signal is determined by the polarity of the baseband signal s(t), independent of its amplitude, thus reducing the power consumption of the system. When s(t) is a three-level coded signal (−1, 0, 1), the dual-band phase-coded pulse signal without background noise can be successfully realized and can be directly used for radar transmission. Compared with the phase-encoded signal of the dual-band continuous wave mode, there is no need for pulse truncation operation before transmission, which simplifies the system and is not affected by background noise.
3. Experiment and results
A proof-of-concept experiment was conducted based on the proposed scheme indicated in Fig. 1. A continuous linearly polarized optical light centered at 1550 nm was injected into a PDM-MZM, which has a 3 dB bandwidth of 20 GHz and a half-wave voltage of 3.5 V. The baseband signal s(t) from the AWG (Tektronix AWG7001A) was sent to one RF port of the MZM1 and the MZM2 after a 3 dB power divider. On the other hand, the other RF port of the MZM1 and MZM2 was driven by RF signals of 10 and 15.5 GHz, respectively. In addition, to suppress the optical carrier modulated by the RF signal, it is necessary to adjust the amplitudes of RF1 and RF2 signals using a commercial electric amplifier (EA) to satisfy β1=β2 = 2.4. A TOBF (Yenista XTA-50) follows the output port of the PDM-MZM to filter out higher-order sidebands.
3.1 Generation of dual-band dual-chirp signals for anti-dispersion transmission
As described in the previous section 2.2, we set the baseband signal from the AWG to be a parabolic microwave signal and divided it into 20 pieces with equal amplitude. Figure 3(a) shows the output optical spectrum of the TOBPF. Obvious double sidebands modulated by RF1 and RF2 can be observed. Under the back-to-back (BTB) conditions, the dual-band dual-chirp signals were generated after photoelectric conversion. Figure 3(b) illustrates the measured electrical spectrum, and the corresponding partially enlarged views are depicted in Figs. 3(c) and 3(d). To enhance the precision and efficiency of the measurement, we configured the resolution bandwidth (RBW) and video bandwidth (VBW) to 100 KHz and 3 MHz, respectively. It can be seen that the generated signals are centered at 10 and 15.5 GHz with the power of 2.3 and −4.2 dBm, respectively, and share the same bandwidth of 100 MHz. Note that the photodetector module (Agilent 11982A lightwave converter) used in our experiment includes a built-in low-noise preamplifier that amplifies the converted electrical signal. This leads to a relatively high peak power value measured on the spectrum analyzer. Figures 3(e) and 3(f) indicate the time-domain waveforms and the corresponding instantaneous frequency-time diagram of the generated dual-chirp signal centered at 10 GHz, while Figs. 3(g) and 3(h) are for 15.5 GHz. Closer inspection of Fig. 3(f) / [Fig. 3(h)] shows an up-chirp signal with a frequency from 10 to 10.05 GHz / [from 15.5 to 15.55 GHz] and a down-chirp signal with a frequency from 10 to 9.95 GHz / [from 15.5 to 15 GHz], indicating that dual-band dual-chirp signals are generated successfully with 100-MHz bandwidth. Figure 4 illustrates the ambiguity function image of the generated waveform in terms of time delay and Doppler frequency. Unlike the conventional single-chirp signal with a knife-edge shape, the figure reveals a clear thumbtack-like shape. Additionally, the contour plot in the inset of Fig. 4 shows that the measured −3 dB full-width at half-maximum (FWHM) and Doppler bandwidth are 7.2 ns and 1.1 MHz, respectively. These results indicate that the resulting dual-chirp signal offers superior range-Doppler resolution in radar applications.
Fig. 3. The measured dual-band dual-chirp signals centered at 10 and 15.5 GHz under BTB conditions. (a) Optical spectrum at the output of the TOBPF, (b) electrical spectrum, (c) and (d) enlarged views of (b), (e) and (g) time-domain waveforms and (f) and (h) instantaneous frequency-time diagrams.
Fig. 4. The ambiguity function for the generated dual-chirp microwave waveform with a 100-MHz bandwidth centered at 10 GHz, Inset: the −3-dB contour map.
To evaluate the characteristics of anti-dispersion transmission of the generated dual-band dual-chirp signals, an optical fiber transmission experiment was carried out. Considering that 15.5 GHz is the power fading point of 15 km SMF, we chose a roll of 15 km SMF for signal transmission. The measured dual-band dual-chirp signals by PD after 15-km SMF transmission were shown in Fig. 5. Figures 5(a) and 5(d) are the electrical spectrum of the measured dual-chirp signals centered at 10 and 15.5 GHz with powers of −3.7 and −10 dBm, respectively, with a 6 dB reduction compared to the aforementioned BTB results. This is mainly caused by the difference in fiber insertion loss and detector response, rather than fiber dispersion. Figures 5(b), 5(c), 5(e) and 5(f) are the time-domain waveforms and instantaneous frequency-time diagrams of the generated dual-chirp signals at 10 and 15.5 GHz, respectively, and both corresponding bandwidths are still 100 MHz.
Fig. 5. The measured dual-band dual-chirp signals centered at 10 and 15.5 GHz after 15-km single-mode fiber transmission. (a), (d) electrical spectrum, (b), (c) time-domain waveforms, (e), (f) instantaneous frequency-time diagrams.
To further confirm the CDIP elimination performance of our proposed system, we replaced the 15 km SMF with a 3 dB optical attenuator for signal transmission, and the experimental results are shown in Fig. 6. Figure 6(a) and 6(d) are the measured electrical spectrums at 10 and 15.5 GHz with powers of −3.8 and −10.3 dBm, respectively, and Figs. 6(b), 6(c), 6(e) and 6(f) are the resulting waveforms and the instantaneous frequency-time diagrams, which are consistent with the results after 15 km SMF transmission, indicating that the generated dual-band dual-chirp signals can achieve anti-dispersion transmission.
Fig. 6. The measured dual-band dual-chirp signals centered at 10 and 15.5 GHz after a 3-dB optical attenuator. (a), (d) electrical spectra, (b), (c) time-domain waveforms, (e), (f) instantaneous frequency-time diagrams.
3.2 Generation of dual-band phase-coded pulse signals
As described in Section 2.3 above, we set the baseband signal to be a three-level phase-encoded signal. The coding sequence was a 13-bit Baker code (−1, −1, −1, −1, −1, + 1, + 1, −1, −1, + 1, −1, + 1, −1) plus 12 bits of 0, and the encoding rate was set to 2 Gb/s. Figure 7(a) depicts the measured optical spectrum output by TOBPF, similar to Fig. 3(a), consisting of the optical carrier modulated by the baseband signal and the ±1st order optical sidebands modulated by RF1 and RF2 signals. After photoelectric detection, dual-band phase-encoded pulse signals were successfully generated. Figure 7(b) is the measured electrical spectrum with center frequencies of 10 and 15.5 GHz, corresponding to powers of 2.5 and −2.5 dBm without background noise, respectively.
Fig. 7. The measured (a) optical spectrum at the output of tunable optical filter; (b) electrical spectrum of the generated dual-band phase-coded pulse signals centered at 10 and 15.5 GHz.
Figure 8(a) and 8(b) are the measured time-domain waveforms of the phase-encoded pulse sequences at 10 and 15.5 GHz with a duration of 100 ns, where the enlarged views of the corresponding single phase-coded pulse are illustrated in Figs. 8(c) and 8(d). Figures 8(e) and 8(f) are the phase information extracted from the waveforms Figs. 8(c) and 8(d) using the Hilbert transform. When the encoded signal changes between s(t) = 1 and s(t)=-1, a clear π shift can be observed, the precise π shift only depends on the polarity of the electrical coding signal. To characterize the pulse compression capability of the signal, autocorrelation calculations were performed on the generated phase-encoded pulse signals at 10 and 15.5 GHz, and the results are shown in Figs. 8(g) and 8(h). Both full width at half maximum (FWHM) are 0.5 ns, the peak-sidelobe suppression ratios are 9.53 and 9.21 dB, and the pulse compression ratios (PCRs) are 13, which are in line with the theoretical results. It is worth noting that the characteristics of the AWG can greatly affect the quality of the generated signal. If the amplitude of the baseband signal output by the AWG is either too large or too small, it will affect the phase modulation depth. Additionally, if the high-frequency characteristics of the AWG output signal are poor, the duration of the final phase-encoded pulse signal at the π transition will increase, leading to a deterioration in signal quality. Moreover, the proposed system can generate a dual-chirp signal in one band and a phase-coded signal in the other band by introducing an additional AWG in the scheme shown in Fig. 1 and generating different baseband signals to separately modulate MZM1 and MZM2. Therefore, the proposed scheme can directly generate multi-band phase-encoded pulse signals without background noise, avoiding the extra pulse truncation operation of continuous-wave phase-encoded signals before emission, which has extremely high application potential in future radar systems.
Fig. 8. The measured 10- and 15.5-GHz phase-coded pulse signals. (a), (b) phase-coded microwave pulse sequences and (c), (d) the enlarged views of the corresponding single phase-coded pulse centered at 10 and 15.5 GHz. (e), (f) the phase information extracted from (c) and (d); (g), (h) the calculated autocorrelation.
4. Conclusion
In conclusion, we have theoretically and experimentally demonstrated a compact photonic scheme to generate multi-format dual-band microwave signals without background noise. Based on the same microwave photonic link, the proposed scheme can not only generate dual-band dual-chirp signals with anti-fiber dispersion transmission, which is suitable for dual-band, high detection accuracy, and one-to-many base station radar systems; but also generate dual-band phase-encoded pulse signals, which can be directly used in the multi-band radar system and does not need extra pulse truncation operation before transmission. In addition, the generated multi-format dual-band signal is free from background noise, avoiding the baseband crosstalk. Therefore, the proposed microwave photonic signal generator can realize the generation of dual-band multi-mode microwave signals with a simple structure, low power consumption and cost, which has great prospects for radar systems.
Funding
Chinese National Key Basic Research Special Fund (2018YFE0201200); Strategic Priority Research Program of Chinese Academy of Sciences (XDB43000000); CAS Project for Young Scientists in Basic Research (YSBR-69); National Natural Science Foundation of China (61835010, 62075210, 62235015).
Acknowledgments
The authors wish to thank the anonymous reviewers for their careful reading and valuable suggestions.
Disclosures
The authors declare no conflicts of interest.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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