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Abstract
In this paper, we present a quasi-optically coupled 600-GHz high-temperature superconducting (HTS) sub-harmonic mixer for communication and sensing applications. The mixer features an innovative double-Y-type slot integrated lens antenna, which can efficiently couple the radio frequency (RF) and local oscillator (LO) signals with a small frequency ratio by exciting the half-wave and full-wave resonant current modes on the slot, respectively. Considering the low impedance characteristics of HTS Josephson junctions, a coplanar-waveguide stepped impedance transformer is utilized for minimizing the mismatching loss. A cascaded filter network is designed to prevent the high-frequency signal leakage at both bands while coupling the intermediate-frequency (IF) signal output efficiently. Based on this antenna design and an established HTS step-edge junction technology, a 600-GHz mixer prototype was designed, fabricated and measured, which was compared with the simulation results. The achieved conversion gain and noise temperature are the best performance specs as reported to date for HTS harmonic mixers at comparable frequencies and operating temperatures.
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1. Introduction
Terahertz (THz) wave has the characteristics of broad bandwidth, short wavelength and mediate penetrability, which serves as one of the major technical means to realize ultrahigh speed communications [1,2] and high-resolution sensing applications [3,4]. However, the wireless propagation attenuation is severe for THz waves, especially in rain or fog, resulting in a serious limitation of communication distance. Therefore, communication or sensing systems are seeking heterodyne receiver frontends with high performance in receiving and down-converting high-frequency signals over longer distances. Superconducting mixers with high conversion gain and low noise temperature have attracted a lot of attention in recent years. Compared with those low-temperature superconducting counterparts [5–9], high-temperature superconducting (HTS) Josephson-junction mixers [10–23] operates at higher temperatures and can be cooled by a small cheaper cryocooler, thus more promising for general wireless applications especially for aerospace scenarios.
According to the operation mode, currently reported HTS THz mixers are generally divided into fundamental mixers [10–15] and harmonic mixers [16–23]. The fundamental mixers usually exhibit better performance at similar operating frequencies and temperatures, but they require expensive local oscillator (LO) sources at higher frequencies and suffer from the difficulty of isolation from the radio frequency (RF) signal. Utilizing a beam splitter one can achieve high isolation, but at the cost of considerable signal loss. In comparison, HTS harmonic mixers can readily isolate those two THz signals through a frequency selective surface (FSS), which brings nearly ignorable losses through good quasi-optical designs. But on the other hand, the harmonic mixer could suffer from degraded conversion and noise performance by using a lower frequency LO source. This can significantly be improved by reducing the harmonic number. To the best of our knowledge, there is still no second harmonic or so-called sub-harmonic HTS THz mixer that has been reported in literature.
One major challenge lies in the design of high-efficiency on-chip antenna and circuit for the sub-harmonic HTS Josephson mixer. Despite that a number of on-chip antennas have been designed to realize the dual-band radiation [24–28], they are not suitable for HTS mixers as the Josephson junction impedance is generally lower than 10 Ω thus causing large mismatching losses. A log-periodic on-chip antenna with ultra-wideband characteristics was used to couple the RF and LO signals at two frequencies [17], but its coupling efficiency is lower than -7.5 dB thus unable to meet the requirement of sub-harmonic HTS mixers. Recently, a THz on-chip antenna that enables high-efficiency dual-band radiation by combining the longer meander twin-slot and shorter straight twin-slot, has been used to design an HTS Josephson-junction fourth-harmonic mixer [22]. Nonetheless, such dual-band radiating structure is not ideal for the case of a small frequency ratio as required by the sub-harmonic mixer. The reason is that the straight twin-slot would produce unwanted parasitic radiations at the LO frequency that significantly deteriorates the radiation patterns. Furthermore, it is much more difficult to achieve sufficient isolation between those two resonant twin-slots for closer RF and LO frequencies. Preventing the THz currents from leaking into the intermediate frequency (IF) port, while preserving the radiation modes of the antenna at both bands, imposes an additional great challenge.
In this paper, we present a 600 GHz HTS sub-harmonic mixer coupled using a double-Y-type slot integrated lens antenna. The major creative contributions are summarized as follows: (1) A high-efficiency dual-band on-chip antenna is designed for coupling the RF and LO signals with a small frequency ratio. The dual-frequency radiation is enabled by exciting the half-wave and full-wave resonant current modes on the slot, and a double-Y-type slot structure is utilized for enhancing the radiation symmetry. A coplanar-waveguide (CPW) stepped impedance transformer is utilized for improving the coupling efficiency for HTS Josephson junctions of low impedance characteristics. (2) To prevent the leakage of THz signal currents, a cascaded CPW filter network is designed with high rejection at both bands. The whole filter has been carefully studied and optimized in such way that, a good virtual ground is formed at the center of the double-Y-type slot for both bands, thus maintaining the radiation modes of the antenna unchanged. Meanwhile, the CPW network transmits the IF signal efficiently. (3) A 600-GHz HTS sub-harmonic mixer prototype is designed, fabricated, and measured. The measured results agree well with the simulated ones using our recently developed modeling method (to be published elsewhere) for HTS harmonic mixers. The best performance specs were demonstrated from the presented HTS mixer prototype in comparison to those reported to date for HTS harmonic mixers at comparable frequencies and operating temperatures.
2. Design and simulations
Considering the operation mechanism of the 600-GHz HTS sub-harmonic mixer, it is necessary to design a dual-band on-chip antenna with high coupling efficiencies at the 300 GHz and 600 GHz for coupling the LO and RF signals, respectively. Figure 1(a) shows the structure and geometry of a double-Y-type slot integrated lens antenna. Being consistent with the YBa2Cu3O7-x (YBCO) HTS material in terms of lattice constants and thermal expansion coefficients, the MgO material is chosen as the substrate of the slot radiator, which has a relative permittivity of 9.63 and thickness of 0.5 mm compatible for device fabrication. Since the surface conductivity of the YBCO HTS film is less than that of the gold film at the frequency above 100 GHz, the top layer that includes the slot antenna and CPW networks is electromagnetically modelled as an infinitely thin gold pattern for reducing the calculation efforts. Taking advantage of different resonant current modes, the double-Y-slot radiator can achieve relatively good radiation symmetry at both the frequencies of 300 and 600 GHz. On the right side of the double-Y-slot, a CPW stepped impedance transformer is utilized for improving impedance matching with the HTS Josephson junction at both bands. A well-designed CPW cascaded filter network is utilized for preventing the leakage of THz currents into the IF signal port. In addition, a hemispherical lens made of the high-resistivity silicon (relative permittivity: 11.28) and with size of 3 mm in diameter, is attached on the back side to achieve high-directional radiation in the negative Z direction and suppress the unwanted surface wave effect caused by the MgO substrate.
Fig. 1. (a) Structure and geometry of a double-Y-type slot integrated lens antenna. (Hsub = 0.5 mm, Rlen = 1.5 mm, Wa = 3 µm, La = 100 µm, L3 = 54 µm, w02 = 7 µm, g02 = 3 µm, L4 = 46 µm, g01 = 2 µm, θ = 90 °, L1 = 68 µm, Lf1 = 100 µm, Lf2 = 50 µm, Lf2 = 50 µm, L2 = 51 µm, w2 = 38 µm, wh2 = 6 µm, g2 = 20 µm, wh1 = 8 µm, w1 = 60 µm, g1 = 30 µm, w01 = 36 µm, and L5 = 3.5 µm). (b) Simulated surface electric currents and electric fields on the double-Y-type slot integrated lens antenna.
Figure 1(b) shows the simulated surface electric current and electric fields on the double-Y-type slot integrated lens antenna, which is obtained by numerical simulations using the software Computer Simulation Technology (CST) Microwave Studio [29]. Clearly, the double-Y-type slot radiator exhibits the desired current modes at the frequencies of 300 GHz and 600 GHz. More specifically, the half-wave and full-wave resonant radiation modes are achieved for the lower and higher operating bands, respectively. Benefitting from the well-designed CPW cascaded filter network, the THz electric currents are effectively isolated from the IF port at the leftmost end. In particular, a good virtual ground is formed at the center of the double-Y-type slot for both bands, thus maintaining the radiation modes of the antenna unchanged. To explain the good radiation characteristics of the antenna, the electric fields along the double-Y-type slot are decomposed into their components in the X and Y directions [see Fig. 1(b)], respectively. In the two principal planes (i.e., the XZ and YZ planes), the equivalent magnetic currents from those electric fields generate radiation fields that cancel out in the X direction while add up in the Y direction, which results in a low cross-polarization level and good linearly-polarized radiation along the Y direction. Moreover, since the double-Y-type slot structure consists of four symmetrically distributed radiation, the equivalent magnetic current sources can approximately be considered as being made of multiple four-element magnetic-dipole square arrays with different side lengths. Hence, good radiation symmetry can be achieved in the XZ and YZ planes.
To better understand the operating mechanism, detailed parametric studies have been carried out to optimize the antenna performance. It is readily understood that the parameters θ and L5 [see Fig. 1(a)] would play dominant role in influencing the radiation symmetry of the antenna. Figures 2(a)–2(b) shows the simulated 3-dB beam widths with the variation of θ and L5 for different operating frequencies. As shown in Fig. 2(a), the 3-dB beam widths in the XZ and YZ planes generally get closer with increasing θ for both frequencies; the best radiation symmetry occurs at θ = 103° for the 600-GHz band. Such phenomenon can be well explained from the simulated electric field distribution as shown in Fig. 1(b). For 600 GHz, the equivalent magnetic currents mainly concentrate at the central part of four tilted slots, the flare angle θ close to 90° is most advantageous for symmetrical radiation. For 300 GHz, since there is still a little magnetic current on the vertical slot, a larger flare angle brings some compensation for symmetry. However, it should be noted that too large θ has detrimental impacts on the CPW network due to the ground damage. As shown in Fig. 2(b), the 3-dB beam widths in the two principal planes turn to be closer with decreasing the length L5. Taking into full consideration the radiation symmetry and CPW transmission characteristics, the parameters θ and L5 are finally optimized to be 90° and 3.5 µm, respectively.
Fig. 2. (a)-(b) The influences of the parameters θ and L5 on the 3-dB antenna beam widths for different operating frequencies, respectively. (c) The influence of g01 on the antenna input impedances at different operating frequencies.
Apart from the radiation characteristics, impedance matching is another important consideration. In absence of the matching network, the double-Y-type slot radiator has a high input impedance of about 320 Ω at 300 GHz and a low impedance of around 19 Ω at 600 GHz, which are determined by the half-wave and full-wave current resonant modes, respectively. To improving impedance matching between the antenna and the HTS Josephson junction (with normal resistance typically in the range of 2-10 Ω at 300 GHz, our first consideration is utilizing a CPW quarter-wavelength line to reduce the high slot impedance, which would not influence the relatively low input impedance at 600 GHz. However, considering RF coupling is significant to the mixer conversion gain, it is necessary to further improve the coupling efficiency at 600 GHz. Here, a CPW stepped impedance transformer has been designed to improve impedance matching and thus coupling efficiency at both bands. Using the stepped impedance transformer as shown in Fig. 1, the input impedance at the antenna port Zin can be written as
Hence, the input impedance at 600 GHz can effectively be reduced by tuning the ratio of those two characteristic impedances. At the frequency of 300 GHz, considering L3 and L4 are equal to around λg / 8 and that Z01 and Z02 are far less than the slot impedance Zslot, Eq. (1b) can be approximated to be
It is seen from Eq. (3) that the high slot impedance at 300 GHz is significantly reduced. Despite that there appears a reactance term in Eq. (3), its value is actually very small according to the following estimations. For the convenience of chip microfabrication, Z01 is set to 50 Ω and the value of Z02 is in the range of 40 Ω < Z02 < Z01. According to Eq. (3), the estimated input resistance at 300 GHz is below 10 Ω and the reactance is in the range of 0-3 Ω. Therefore, the input impedance at both frequencies can be reduced by using the CPW stepped impedance transformer. Figure 2(c) shows the simulated input impedance with the variation of g01 for different operating frequencies, which is obtained by using the CST software. Clearly, the input resistance at both bands drops with the decrease of Z02, which is consistent with the conclusion as predicted by Eqs. (2) and (3). Due to some parasitic effects, there is minor variation for the input reactance varies but the value keeps in the vicinity of zero. Taking into consideration the manufacturing limitation and that the input impedance should be sufficiently low for high coupling efficiency, the gap g01 is finally chosen to be 2 µm.
For heterodyne mixing application, since the IF signal is coupled from the HTS Josephson junction into the readout circuit, a choke filter network is required for preventing the leakage of the THz electric currents. As shown in Fig. 1, a CPW cascaded filter network has been designed with high rejection at both the 300-GHz and 600-GHz bands. Figure 3(a) shows the simulated reflection and transmission coefficients of the CPW cascaded filter network. For both frequencies, the reflection coefficients S11 are above -1.5 dB and the transmission coefficients S21 are below -20 dB, which shows the good isolation characteristics of the cascaded filter. It should be mentioned that the presence of such filter network might alter the surface current modes that would deteriorate the radiation performance of the antenna. To avoid deterioration, a virtual ground must be formed at the center of the double-Y-type slot for both bands. Accordingly, the input impedance of the CPW cascaded filter network has been carefully investigated by single-port simulation. Figures 3(b)–3(c) show the simulated input impedance at port 1 with the variation of L2 and L1, where the port 2 is connected to a 50-Ω matched load. It is seen that due to the well isolation of the higher-band chock filter, the variation of L1 has very little influence on the impedance characteristics at the 600-GHz band. In comparison, the variation of L2 can tuning the input impedance at both bands. Hence, the optimization procedure starts with adjusting L2 to make the impedance approach zero at 600 GHz, and then changes L1 to perform impedance tuning at 300 GHz. Finally, it can be found that the virtual ground condition can approximately be satisfied by using the parameters of L2 = 51 µm and L1 = 68 µm.
Fig. 3. (a) The reflection and transmission coefficients of the cascaded filter network. (b)-(c) The influences of L2 and L1 on the input impedance of the cascaded filter network with port 2 connected to a matched terminal, respectively.
After careful design, analysis and optimization, the radiation performance of the double-Y-type slot integrated lens antenna has been finalized. Figure 4(a) shows the simulated input impedance versus operating frequency. Clearly, the input resistance Rin is 5 Ω at 300 GHz and 12 Ω at 600 GHz, and the input reactance Xin is close to 0 Ω at both bands. Those impedance values are close to that of the HTS Josephson junction, which minimize the mismatching loss thus leading to high coupling efficiencies. Figures 4(b) shows the reflection coefficient S11 and coupling efficiency $\eta_c=(1\hbox{-}|S_{11}|^{2})\eta$, where $\eta$ is the radiation efficiency. It is seen that $\eta_c$ achieves maximum at the frequencies of 300 GHz and 600 GHz, which are -1.9 dB and -2.4 dB, respectively. The 1.5-dB available operating bandwidth is about 40 GHz and 41 GHz, respectively. It should be mentioned that the ripples in the simulated curves are caused by the electromagnetic wave reflection at the interface between the lens and air. Figures 4(c)–4(d) show the simulated co-polarized and cross-polarized radiation patterns at 300 GHz and 600 GHz, respectively. Clearly, the radiation patterns exhibit high directivity toward the negative Z direction and very good symmetry in the XZ and YZ planes. As readily understood from the fixed aperture size of the lens, the 3-dB beam-width reduces from 14.8° at 300 GHz to 10.9 ° at 600 GHz. Besides, the realized gain is 18.7 dBi and 21.5 dBi for the lower and higher operating frequency bands, respectively. The side-lobe and back-lobe levels are at least 14 dB lower than that of the main lobes for both bands. In addition, the cross-polarization levels are lower than -24.7 dBi at 300 GHz and -31.5 dBi at 600 GHz, respectively. All those results have shown the good radiation performance of the presented antenna.
Fig. 4. (a)-(b) Finalized input impedance, coupling efficiency and reflection coefficient of the double-Y-type slot integrated lens antenna. (c)-(d) Finalized radiation patterns of the antenna at 300 GHz and 600 GHz, respectively.
Figure 5(a) shows the finalized IF reflection and transmission coefficients of the CPW thin-film circuit in the frequency range of 1-40 GHz. It should be noted that at the left side of the cascaded filter network, a 50-Ω gradient CPW line is utilized to increase the width of the central conductor thus facilitating the readout and measurement of the IF signal. It is seen that the transmission coefficient S12 or S21 is above -0.6 dB, and the reflection coefficients S11 and S22 are below -15 dB over the whole band of interest. The results have shown the high coupling efficiency and larger IF bandwidth, which is advantageous for the design of wideband HTS THz sub-harmonic mixer. Lastly, a dc biasing circuit is designed to offer dc biasing for the HTS Josephson junction to perform harmonic mixing. The illustration in Fig. 5(b) shows a three-port bias-tee network, which is implemented on a 0.254-mm thick alumina substrate with the dimensions of 10 mm × 5 mm. The bias-tee network provides adequate isolation between IF coupling and dc biasing, which is realized by using a 100-nF capacitor and two 500-Ω resistors. It can block the dc signal from the IF port (port 1), and prevent the IF signal from leaking into the dc port (right pads). Two horizontal dc lines and square pads are used to apply the high-precision four-point testing method. It should be mentioned that the port 1 is connected to a coaxial connector on a packaged house, and the port 2 is connected to the CPW thin-film circuit via gold bonding. Figure 5(b) shows the finalized reflection and transmission coefficients of the dc biasing microstrip circuit in the frequency range of 1-40 GHz. Clearly, the transmission coefficients are higher than -1.2 dB and the reflection coefficients are lower than -13.5 dB, respectively. Hence, the dc biasing circuit can provide high-efficiency IF coupling while offering dc biasing.
Fig. 5. (a) Finalized IF reflection and transmission coefficients of the (a) CPW thin-film circuit and (b) dc biasing microstrip circuit (C1 = 100 nF, R1 = R2 = 500 Ω) in the frequency range of 1-40 GHz.
3. Experimental details
Based on the electromagnetic design and analysis as detailed in Section II, a 600-GHz HTS sub-harmonic mixer was designed and implemented. The double-Y-type slot integrated lens antenna-coupled HTS mixer was fabricated at CSIRO laboratory using the advanced YBa2Cu3O7-x (YBCO) HTS step-edge junction technology [30,31]. Figure 6(a) shows a micrograph of the fabricated HTS mixer device chip. The double-Y-type slot radiator as well as the CPW matching and isolation networks, were patterned on a 300-nm thick gold thin film (the yellow colored part) deposited on a 0.5-mm thick MgO substrate. As shown in the right part of Fig. 6(a), a HTS step-edge junction (the black-colored tiny section) was manufactured at the feeding port of the on-chip antenna where a 2-µm-wide YBCO strip across the edge of a step pattern created on the MgO substrate. The YBCO step-edge junction and the gold thin-film circuit were fabricated by using standard photolithography and ion beam etching techniques [30,31]. The YBCO lines underneath the gold film are dc biasing lines used for biasing and measuring the dc current-voltage characteristic of the HTS Josephson junction.
Fig. 6. (a) Micrograph of the fabricated mixer device chip showing the double-Y-type slot, CPW coupling networks and HTS step-edge junction. (b) The measurement setup for the HTS THz sub-harmonic mixer with a close-up view of the packaged antenna-coupled mixer module.
To reduce the fabrication cost, two mixer devices were implemented on a 10 × 10 mm2 MgO chip with two hemispherical silicon lenses mounted on the backside of the chip, which were then packaged into a specially designed metal house as a HTS sub-harmonic mixer module [see the inset of Fig. 6(b)]. Figure 6(b) also shows the photograph of the experimental setup for characterizing the HTS mixer performance. Generated by two VDI solid-state sources, the RF and LO signals were collimated by an Edmund Optics off-axis parabolic mirror and a Teflon planoconvex lens, respectively. Those two signals were combined via a Tydex beam splitter and then refocused onto the silicon lens of the HTS mixer module by another off-axis parabolic mirror. A temperature adjustable pulse tube cryocooler [32] was utilized for cooling the HTS mixer, which has a window for coupling the RF and LO signals. To keep the mixer device operating within its linear region, a rotatable wire-grid polarizer that functions as an attenuator was applied in the link to make RF signal power sufficiently lower than the LO power. A battery-operated dc current source was used to bias the HTS Josephson junction. The down-converted IF signal from heterodyne mixing, was amplified by a low-noise amplifier (LNA) with gain of 26-30 dB over 4-24 GHz, and then recorded using a Keysight E4407B spectrum analyzer.
To start with mixer performance characterizations, the IV characteristics of the HTS Josephson junction was measured to obtain the junction parameters, i.e. the junction critical current Ic and normal resistance Rn, and decide the optimal bias point. The mixer conversion gain was obtained by using the formula Gmix = PIF / PRF / GIF, where PIF is the IF output power measured using the spectrum analyzer, GIF is the calibrated IF link gain, and PRF is the power of the RF input signal. PRF can be obtained from the coupled power into the Josephson junction by measuring the Ic suppression and Shapiro step heights from the pumped IV curves. Considering the complexity and inconvenience in building the hot/cold-load experimental setup for noise characterization at such high frequencies and with quasi-optical coupling scheme, an alternative approach instead of the Y-factor method was used due to its conveniences and enough precision for harmonic mixers. By replacing the RF source with a THz blackbody made of an absorber, we recorded the noise power Pout at the IF output port under the optimal dc biasing and LO pumping conditions. Then, the mixer noise temperature Tmix can be obtained according to Eq. (4),
4. Measured and simulation results
To the best of our knowledge, there still lacks accurate and efficient modeling method for analyzing the noise and conversion properties of HTS harmonic mixers. One major difficulty is that HTS Josephson-junction harmonic mixers generally feature much more complicated spectrum than fundamental ones. Another challenge is that HTS harmonic mixers have chaotic behaviors as determined by the superconducting Josephson equation [34], making it unrealistic to analyze all or most parasitic components with reasonable computation complexity. To tackle this issue, we have recently presented a parasitics-selection based equivalent circuit model by quantitively comparing and analyzing the conversion efficiency contribution of the RF and parasitic components, which significantly reduces the complexity to a six-port network thus easing the analysis pressure. To calculate the conversion impedance matrix of the model, we present a fast hybrid-solution method that combines the single-tone current excitation and perturbation-derivation processing techniques. Detailed derivations are then carried out to acquire the semi-analytic expressions of the noise temperature and conversion gain. The modeling method has been coded using MATLAB and its effectiveness verified by experiments, for which the specific details are to be published elsewhere. Here, we apply the modeling method to analyzing the performance of the presented HTS harmonic mixer in this work.
Figure 7 shows the measured and simulated dc IV characteristics at the operating temperatures of 40 K, 60 K and 70 K, respectively. The unpumped dc IV curves, as shown in Fig. 7(a), display the typical resistively-shunted-junction behavior; the normal resistance Rn is about 2.1 Ω and the critical current Ic is 925 μA at 40 K, 430 μA at 60 K and 220 µA at 70 K, respectively. While the LO and RF signals are coupled into the Josephson junction by the lens antenna, the Ic is suppressed to some degree and a series of periodic Shapiro steps appear on the IV curves, as shown in Figs. 7(b)–7(c). In experiment, the LO and RF frequencies were selected to be 290.64 GHz and 590.4 GHz, respectively, where relatively high output power was generated by the THz sources. Clearly, the voltage period of the Shapiro steps induced by LO pumping is near half of those for RF pumping, which complies well with the Josephson frequency-voltage relationship [35]. The LO and RF pumping power (i.e., the PLO and PRF) are characterized to be around -35.8 dBm, and both they were measured in the same way. As clearly seen from Fig. 7, all measured IV curves are in good agreement with the simulated results and very minor differences could be due to minor fabrication defects and experimental errors.
Fig. 7. Measured and simulated dc IVCs of the HTS THz sub-harmonic mixer for different operating temperatures when (a) unpumped, (b) pumped by a LO signal and (c) pumped by a RF signal.
Figure 8(a) shows the measured and simulated conversion gain Gmix versus the dc bias current IB at the temperature of 40 K. To have the mixer operate within its linear region, the RF signal power PRF was reduced to be -54.5 dBm by used the THz attenuator, which is 18.7 dB lower than the LO pumping power PLO. It can be clearly seen that, the Gmix versus IB curves exhibits obvious fluctuation characteristics with several peaks. As can be readily understood from the dc IV characteristics under LO pumping [see Fig. 7(b)], those peaks occur in the middle of the adjacent Shapiro steps where the dynamic resistance RD (i.e. the slope of the IV curve) reaches its maximum value. The measured result exhibits a double-peak behavior between the second and third Shapiro steps, which is different from the simulated one with one peak only. The underlying reason is still unclear and further study is required to understand this behavior. On the whole, the measured and simulated results agree reasonably well in terms of the curve shape, peak positions and gain values. The first two peaks agree particularly well. The best measured and simulated Gmix are around -17.5 dB and -15.2 dB, respectively.
Fig. 8. (a) Measured and simulated mixer conversion gain Gmix versus the dc biasing current IB at 40 K. (b) Measured and simulated mixer noise temperature Tmix versus the dc biasing voltage Vdc at 40 K. (c) Measured and simulated mixer noise and conversion gain versus the LO power PLO at different temperatures.
Figure 8(b) shows the noise temperature Tmix of the HTS THz sub-harmonic mixer versus the dc biasing voltage Vdc at the temperature of 40 K. Obviously, the mixer noise temperature Tmix is dependent on the biasing voltage Vdc, which degrades sharply at the edges of the Shapiro steps. This is mainly caused by the discontinuity of the pumped dc IV characteristics at these locations [see Fig. 7(b)], where the harmonics of the LO signal interfere with the Josephson oscillations. Hence, the HTS mixer performs well when biased in the flat region which is around halfway between the Shapiro steps on the IV curve. The mixer noise performance was characterized using the method as detailed in Section A. At optimal dc biasing and LO pumping conditions, the noise temperature Tmix of the HTS mixer module was measured to be around 7670 K, which agrees well with that from numerical simulation.
Figure 8(c) shows the mixer noise temperature Tmix and conversion gain Gmix versus the LO power PLO at different temperatures. Clearly, the Tmix and Gmix values vary more dramatically for PLO less than -38 dBm and tend to flatten with further increasing PLO value. This is mainly because that the pumping power of smaller LO signal and thus its second harmonic are too weak to establish stable heterodyne mixing in the HTS Josephson junction. As observed from the simulated results, the best possible conversion gain and noise temperature are -15 dB and 4130 K at 40 K for current junction parameters with relatively low Rn and high Ic, respectively. Provided using better junction parameters as measured from our previously manufactured HTS junction [15], the mixer performance can be significantly improved for identical antenna and circuit designs as presented in this work. As clearly shown in Fig. 8(c), the Tmix and Gmix for new junction parameters achieve around -13 dB and 1650 K when operating at the temperature of 20 K, respectively. Due to the very high critical current, it is difficult to characterize the parameters of current Josephson junction at 20 K and thus corresponding mixer performance. At a fixed LO power of -35.85 dBm, the measured Tmix and Gmix results at 40 K were also plotted in Fig. 8 for comparison. Although the measured device performance is less optimal compared to the simulated one, it agrees reasonably well with the simulated one considering the likely fabrication imperfection, measurement and calibration errors in the process.
Table 1 compares the performance of the presented HTS sub-harmonic mixer with other state-of-the-art superconducting harmonic mixers in the same frequency range. The device implementation technologies include the superconductor-insulator-superconductor (SIS) tunnel junction and YBCO step-edge junction technologies. As one major advantage of superconducting mixers, the LO pumping power is relatively low for all listed harmonic mixers. It shows that our presented mixer exhibits superior HTS device performance in terms of conversion gain and noise temperature at the comparable operating temperatures. It should be emphasized that if having the parameters of our previously developed Josephson junction, the mixer performance can be further improved for identical device designs as presented in this work, especially when operating at 20 K. We attribute the superior performance to our innovative work that includes creative on-chip antenna designs, advanced matching and isolation networks, and effective modeling simulation analyses.
Table 1. Performance comparison of state-of-the-art superconducting harmonic mixers in the 600 GHz band*
5. Conclusion
In this paper, a 600-GHz HTS Josephson-junction sub-harmonic mixer has been presented for communication and sensing applications. An innovative double-Y-type slot integrated lens antenna is designed to couple the RF and LO signals into the mixer with high efficiency and a small frequency ratio. Good radiation performance has been achieved by extensively investigating the resonant current and field modes on the slot radiator, as well as utilizing a coplanar-waveguide stepped impedance transformer for improving the matching with the low-impedance HTS Josephson junction. A cascaded filter network is also well designed to prevent the THz signal leakage while not influencing IF coupling. On the basis of the electromagnetic designs, a 600-GHz HTS sub-harmonic mixer prototype was designed, fabricated and measured. For comparison, an innovative HTS harmonic mixing modeling method is applied to analyzing the device performance. The achieved conversion gain and noise temperature are the best performance specs reported to date for HTS harmonic mixers at comparable frequencies and operating temperatures. Considering the merits of high sensitivity, low LO power and small cheaper cooling system, the presented HTS mixer will find great potential in THz communication and sensing applications.
Funding
National Key Research and Development Program of China (2019YFB1803200); National Natural Science Foundation of China (62171032, 61971038); CSIRO Space Future Science Program Project (ST-R2-03).
Acknowledgments
The authors would like to thank Ms. Jeina Lazar in CSIRO Manufacturing for the HTS chip fabrication.
Disclosures
The authors declare that there are no conflicts of interest related to this article.
Data availability
Data underlying the results presented in this paper are not publicly available at this time but may be obtained from the authors upon reasonable request.
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