Reconfigurable multi-band microwave photonic radar transmitter with a wide operating frequency range

Xiangpeng Zhang; Xiangpeng Zhang; Qiang Sun; Qiang Sun; Jiyao Yang; Jiyao Yang; Jiming Cao; Jiming Cao; Wangzhe Li; Wangzhe Li

doi:10.1364/OE.27.034519

1. Introduction

Implementation of multi-function or multi-band radar transmitter in a unified hardware structure contributes to better target detection, imaging and cognition [1–3]. However, for a conventional electronic radar transmitter, one function or one band is usually associated with a set of hardware. To achieve multi-function or multi-band operation, more than one set of hardware is required and the complexity of the system would significantly increase.

To overcome the difficulties of electronic counterparts, photonics-assisted multi-band or multi-function radar transmitters have been proposed, owing to photonics’ inherent advantages, such as low transmission loss, immunity to electromagnetic interference and ultra-wide bandwidth [4–7]. The world’s first photonics-assisted multi-band radar transmitter is proposed by Prof. Bogoni’s group [8]. Multiple phase-locked optical carriers produced by a passive mode-lock laser (MLL) are modulated respectively by different signals. Each modulated carrier is then converted into microwave signal in different frequency band. Signals in S band and X band are selected by two bandpass filters with corresponding central frequencies. The similar structure is utilized to realize a dual-use radar transmitter to achieve communication and radar detection [9]. However, the bandwidths of the generated signals are several megahertz suffering from the limited repetition rate of the passive MLL. A multi-band radar transmitter based on an integrated polarization multiplexing dual-parallel Mach-Zehnder modulator (MZM) is demonstrated by Prof. Pan’s group [10]. Both bandwidth and central frequency are quadrupled compared with the input RF signal, contributing to a high range resolution in radar imaging [11–15]. However, the bias voltage of the modulator and the amplitude of the driving signal have to be precisely adjusted to achieve excellent harmonic suppression. Due to either bandwidth limitation or operation difficulty, the aforementioned microwave photonic multi-band radar transmitters are suitable for only a few specific applications.

In this letter, a novel approach to achieve a reconfigurable photonics-based multi-band radar transmitter is proposed. A tunable OEO combined with a microwave photonic frequency-converting link realizes the generation of multi-band signal. The scheme offers three significant advantages. Firstly, owing to broad frequency tunable range of the OEO and a software-defined RF source, the central frequency, bandwidth and temporal period of the generated transmission signals can be programmable. Secondly, multi-band radar transmitters based on commercial RF sources suffer from bandwidth limitation and phase noises in high frequencies. The OEO-based scheme produces multi-band transmission signals with large frequency ranges, ultra-low phase noises and multiplied bandwidths, reducing the waveform distortion and improving the system’s dynamic range [16]. Thirdly, the proposed multi-band transmitter can achieve six band operations and all the multi-band signals can be coherent using a phase-locked loop, implementing more functions with higher precision and resolution [17]. An experiment in the detection of a moving target is conducted. The target’s velocity and range information are obtained, which validate the performances of the developed multi-band radar transmitter.

2. Principle of operation

The schematic of the microwave photonic multi-band radar transmitter is depicted in Fig. 1. The scheme includes a tunable OEO, a low-frequency RF source and a microwave photonic frequency-converting link. The key component in the structure is the tunable OEO, which is employed to generate a single tone with ultra-low phase noise. A narrow-band microwave waveform provided by the RF source is mixed and frequency converted with the single tone in the microwave photonic frequency-converting part to generate multi-band transmission signals. The RF-chains before the transmitting antenna consist of electronic devices, such as electrical amplifiers and filters, outputting amplified signals in desired frequency bands. The operation principle of the proposed radar transmitter focuses on the tunable OEO and the microwave photonic frequency-converting link.

Fig. 1. Schematic of the microwave photonic multi-band radar transmitter; CW: continuous-wave light source; PolM: polarization modulator; OC: optical coupler; PC: polarization controller; MZM: Mach-Zehnder modulator; PM: phase modulator; TBPF: optical tunable bandpass filter; LNA: low noise amplifier; PA: power amplifier; PS: phase shifter; ATT: attenuator; EDFA: erbium-doped fiber amplifier; Div: power divider; PD: photodetector; RF SG: RF signal generator.

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2.1 Triple-loop OEO

As shown in Fig. 1, the setup of the proposed triple-loop OEO contains a continuous-wave (CW) light source, a polarization modulator (PolM), an optical coupler (OC), two polarization controllers (PC1 and PC2), two phase modulators (PM1 and PM2), two spools of single mode fibers (SMF1 and SMF2) with different lengths, a polarization-maintaining fiber (PMF), an optical tunable bandpass filter (TBPF), a photodetector (PD1), an erbium-doped fiber amplifier (EDFA1), a low noise amplifier (LNA), a power amplifier (PA), an attenuator (ATT) and two power dividers (Div1 and Div2).

An incident light wave from a CW light source is sent to a PolM via PC1 which makes the polarization direction of the light wave have an angle of 45° with respect to one principal axis of the PolM. The PolM is capable of enabling phase modulation at two orthogonally polarized principal axes with opposite modulation indices [18,19]. By properly controlling PC2, one polarization axis of the PolM is aligned with the polarization axis of PM1. The PolM is then equivalent to a PM. The light wave is phase modulated in the PolM by a microwave signal excited from the configuration noises and different orders of sidebands are generated. The maximum power of Div1 output signal is set as 11 dBm, and the half-wave voltage of the modulators is around 7 V, which means the modulation index is less than 0.49. Under small signal modulation, only carrier and ±1^st-order sidebands are considered. If the modulated light wave is directly sent into PD1, no RF signal is detected due to the phase modulation. However, if one sideband is suppressed, a phase-modulation-to-intensity-modulation (PM-IM) conversion can be achieved and the microwave signal can be recovered at the output of PD1. The PM-IM conversion has a transfer function equivalent to a microwave photonic filter (MPF). The passband of the MPF in the OEO cavity is determined by the difference between the frequency of the optical carrier and the central frequency of the TBPF. The signal from the MPF is amplified by the amplifiers and split into two parts by Div1. One signal is sent to the PolM to form a close loop and the other signal is separated by Div2 into two paths and respectively applied to different PMs.

A long fiber loop is applied in the developed OEO to reduce the phase noise of the oscillation signal. However, finding an optical filter narrow enough to sustain a single-mode operation is difficult due to that the mode spacing is inversely proportional to the feedback loop length [20]. To facilitate the single-mode selection, a triple-loop technique is employed in the OEO structure. Benefiting from the Vernier effect, only if the phase delays of the signal are multiples of 2π after each loop circulation, the signal with corresponding frequency starts oscillation. This method eases the bandwidth requirement of the TBPF and makes the oscillator widely tunable and practically stable [21,22]. More details on the operation principle can be found in [23].

2.2 Microwave photonic frequency-converting link

The schematic of the microwave photonic frequency-converting portion is sketched in Fig. 2, which consists of a PC3, a polarizer, an MZM, an RF signal generator, an EDFA2, and a PD2.

Fig. 2. Schematic of the microwave photonic frequency-converting link; A, B (red): Optical spectra at different locations in the microwave photonic link; C, D (black): Electrical spectra at different locations in the microwave photonic link.

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Firstly, the incident light wave, a pair of complementary phase-modulated signals from the PolM, is sent to the polarizer via PC3. The polarizer is aligned in 45° with one principal axis of the PolM. By controlling PC3, a static phase difference (π) between the light carriers with orthogonal states of polarization can be added. A double-sideband with carrier suppressed (CS-DSB) modulation is thus completed, as shown at point A. Secondly, the output light wave from the polarizer is modulated by a narrow-band RF signal (shown at point C) in the MZM biased at the minimum transmission point (MITP). Finally, the modulated light wave (shown at point B) is amplified by EDFA2 and fed to PD2, where a frequency-multiplied signal, a frequency-up-converted signal and a frequency-down-converted signal are generated simultaneously (shown at point D).

Mathematically, assuming the amplitude and angular frequency of the optical carrier are E_c and ω_c, respectively. The light wave after the polarizer can be expressed as

(1)$${E_{PA}}(t) \approx {E_1}\exp (j{\omega _\textrm{c}}t + j\frac{{\pi }}{2}) \times \cos ({\omega _e}t)$$

where E₁=4E_cJ₁(πV_e/V_π1), V_e and ω_e denote the amplitude and angular frequency of the oscillation signal, respectively. J₁ is the first-order Bessel function of the first kind. V_π1 is the half-wave voltage of the PolM. As can be seen from Eq. (1), the carrier (0th-order Bessel function) is suppressed. The MZM is also biased at the MITP, the light wave at the output of the MZM can then be directly written as

(2)$${E_{\textrm{MZM}}}(t) \approx {E_\textrm{2}}\exp (j{\omega _c}t) \times \cos ({\omega _e}t) \times \cos ({\omega _0}t + \pi k{t^2}),\textrm{ }T/2 < t < T/2$$

where E₂=−4E₁J₁(πV₀/V_π2), V₀ and ω₀ are the amplitude and the central angular frequency of the linearly frequency-modulated continuous wave (LFMCW) signal produced by the RF signal generator, respectively. k is the positive chirp rate; T is the temporal period of the LFMCW signal and V_π2 is the half-wave voltage of the MZM. When the dc component is ignored, the output of the PD2 can be expressed as

(3)$$\begin{array}{c} {V_{\textrm{PD}}}(t) \propto \cos [{2{\omega_{\mathop{\textrm {e}}\nolimits} }t - \textrm{2(}{\omega_0}t + \pi k{t^2}\textrm{)}} ]+ \textrm{2}\cos [{2\textrm{(}{\omega_0}t + \pi k{t^2}\textrm{)}} ]\\ + \cos [{2{\omega_e}t + \textrm{2(}{\omega_0}t + \pi k{t^2}\textrm{)}} ]+ \textrm{2}\cos (2{\omega _e}t) \end{array}$$

the first three terms on the right-hand side of Eq. (3) correspond to three output waveforms in different bands. The single tone can be filtered out by the RF chains. As also can be seen from Eq. (3), the power of the frequency-multiplying signal is theoretically 6 dB larger than that of two others.

The bandwidth of the generated waveforms is twice that of the input RF signal, contributing to a better detection precision. Besides, the single tone is provided by the tunable OEO, which indicates that the central frequency of the waveforms generated via microwave photonic mixing can be controllable. However, the frequency constraint condition of avoiding spectral overlap between different bands are considered and expressed as

(4)$$\begin{array}{l} {\omega _e} - {\omega _0} + \pi kT/2 < {\omega _0} - \pi kT/2\\ {\omega _0} + \pi kT/2 < {\omega _e}\\ {\omega _e} < {\omega _e} + {\omega _0} - \pi kT/2 \end{array}$$

the condition can be simplified as

(5)$$\begin{array}{c} {\omega _0} + {\pi }kT/2 < {\omega _e} < \textrm{2}{\omega _0} - {\pi }kT\\ {\omega _0} > \pi kT/2 \end{array}$$

According to Eq. (5), to obtain signals in a desired band without inter-band interference, the parameters of the LFMCW signal and the oscillation signal should be appropriately selected.

3. Experiments and results

Based on the setup shown in Fig. 1, a proof-of-concept experiment is performed. Upon arrival at the PolM (Versawave), a light wave with a high power of 19 dBm from a CW laser (TeraXion) is equally split into two paths. The light wave along one path is fed into the OEO section; the light wave in the other path is injected into the microwave photonic frequency-converting link. In the OEO section, the lengths of delay lines (SMF1 and SMF2) are 10 m and 4 km, respectively. A PMF with a length of 450 m provides a third time delay and mitigates the polarization mode dispersion of the fibers between phase modulators. A phase-shifted fiber Bragg grating (PS-FBG) with a 40-GHz central frequency tunable and 110-MHz passband is employed as a TBPF. The signal output from PD1 is amplified by electrical amplifiers with the total gain of around 55 dB. When the triple loops are closed, a fundamental microwave oscillation signal is generated.

In the microwave photonic frequency-converting link, by adjusting PC3, the incident light wave is CS-DSB modulated by the oscillation signal. When the frequency of the oscillation signal is set as 12.4 GHz and the frequency range of the RF source (Tektronix 70001A) is set from 7 GHz to 7.8 GHz, the optical spectra of the light wave are measured by an optical spectral analyzer (Yokogawa AQ6370D) with a resolution of 0.02 nm. In Fig. 3(a), the measured frequency difference between ±1st-order sidebands is 24.75 GHz. The power suppression ratio between the carrier and the ±1st-order sidebands is 22.12 dB, which is of great importance to implement photonic mixing with negligible spurs. The optical spectrum of the MZM light wave is shown in Fig. 3(b). The optical sideband suppression ratio is 19.22 dB.

Fig. 3. (a) Optical spectrum of light wave after being CS-DSB modulated by the oscillation signal; (b) optical spectrum of the light wave at output of the MZM.

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The characteristics in terms of frequency tunable and phase noise of the OEO are evaluated by an electrical spectrum analyzer (ESA) and a signal source analyzer (Agilent Technologies E5052B), respectively. Figure 4(a) shows that the oscillation frequency is tuned from 6.5 to 15 GHz with a tuning step of 500 MHz. The frequency tunable of the oscillation signal is limited to the bandwidths of the PS-FBG and the RF transmission lines but sufficient to meet the requirements for multiple functions. The RF gain fluctuations mainly result from the nonlinearity of the phase/polarization modulators. Due to the nonlinearity of modulators and the electrical amplifiers, the second-order harmonic wave of the desired signal also can be seen in the spectra. Other extra signals whose frequency is very close to the oscillation signals result from of the sidemodes of the OEO. The single-sideband phase noise spectrum of the experimentally generated 13-GHz microwave signal is shown in Fig. 4(b). The phase noise at 10 kHz away from the carrier is –127.1 dBc/Hz. The peaks which have a frequency spacing corresponding to a free spectral range of the OEO result from the non-oscillating sidemodes. A phase noise performance of a commercial RF source (Agilent Technologies E8267D) and the triple-loop OEO is also compared in Fig. 4(b), showing the superiority of phase noise performance of the OEO. The green curve in Fig. 4(b) is a simulation result with the fiber length ratios consistent with the experiment using a model proposed in [24]. At a frequency offset below 100 kHz, the measurements are consistent with the simulation result. The phase noise deterioration above the 100-kHz frequency offset in measurements is attributed to OEO cavity’s additive noises, including thermal noise, high-frequency RIN induced noise and shot noise. The stability of the OEO configuration is also measured and a little worse than that of the OEO proposed in [23]. Using the same model, the optimal phase noise performance is obtained by setting the delay ratios as 50:5:1 among the triple loops and the spurs are suppressed effectively, as shown in Fig. 4(b) (bold purple curve). Choosing lasers with low RIN level and adding thermal control can further improve phase noise performance and stability of the oscillation signal.

Fig. 4. (a) Measured oscillation frequency from the tunable OEO; (b) phase noise comparison among an RF source (in blue curve), a triple-loop OEO (in red curve), and two simulation results with the actual (in green curve) and ideal (in purple bold curve) fiber length when generating signals with frequency of 13 GHz.

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When the OEO is stable, the generations of multi-band transmission waveforms in six frequency bands are implemented. As can be seen from Eq. (3), the transmission signal generated by microwave photonic frequency-multiplying only depends on the RF signal. By modifying the frequency range of the input RF signal as 8.1–8.9 GHz, 12.6–13.4 GHz and 14.1–14.9 GHz, transmission signals in Ku (16.2–17.8 GHz), K (25.2–26.8 GHz) and Ka (28.2–29.8 GHz) frequency bands are generated, respectively. When the frequency of the input RF signal is set from 8.1 to 8.9 GHz and the OEO’s oscillation frequency is set as 10 GHz, S-band frequency-down-converted signal (2.2–3.8 GHz), Ku-band frequency-multiplied signal (16.2–17.8 GHz) and Ka-band frequency-up-converted signal (36.2–37.8 GHz) are generated simultaneously. After that, keep the RF signal’s frequency fixed and set the frequency of the OEO as 12 GHz, C-band signal (6.2–7.8 GHz), Ku-band signal (16.2–17.8 GHz) and U-band signal (40.2–41.8 GHz) are generated at the same time. When we set the OEO’s frequency as 13.8 GHz and keep the RF signal unchanged, X-band signal (9.8–11.4 GHz), Ku-band signal (16.2–17.8 GHz) and U-band signal (43.8–45.4 GHz) are generated at the same time. The generated signals in six bands are shown in Fig. 5, while, the signals with frequency higher than 35 GHz are not recorded due to the lack of the electrical filters in such high frequencies. The in-band power variations of the produced waveforms are mainly caused by the nonlinearity of the modulators and the standing waves induced by RF connectors impedance mismatch. The bandwidth of the homemade electrical filters in each band is 2 GHz to suppress out-of-band spurious signals, which limits the tunability of the scheme though.

Fig. 5. Spectra of the generated signals in six frequency bands.

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Due to 20-GHz bandwidth limitation of the oscilloscope (Keysight Technologies 92004A, OSC), only transmission signals in four frequency bands (S, C, X and Ku) can be sampled and recorded by the OSC in the time domain. For simplicity, qualities of transmission signals in X and Ku band are investigated. The gain flatness (GF), in-band distortion (IBD) and signal-to-noise ratio (SNR) of the signals are evaluated. In Fig. 5, the GF of the generated transmission signal in X band (9.8–11.4 GHz) is 6.31 dB and the SNR of the generated transmission signal in X band is 32.91 dB. The GF and the SNR of the generated transmission signal in Ku band (16.2–17.8 GHz) is 2.58 dB and 37.63 dB, respectively. The measured X-band and Ku-band waveforms in the time domain are shown in Figs. 6(a) and 6(d), respectively. The power fluctuations are due to the impedance mismatch of the RF components and unflatten frequency transfer responses of the components in the link. The IBDs of the produced signals have been investigated. Figures 6(b) and 6(e) show the generated signals in the frequency-time diagram. As can been seen that in-band spurious are observed in both figures. Typical vertical-sections of the instantaneous frequency-time diagram of the generated signals are provided, shown in Figs. 6(c) and 6(f), respectively. The IBD suppression of the Ku-band signal is 52.14 dB while the measured IBD suppression ratio of the X-band signal is 41.9 dB.

Fig. 6. (a) Waveform of the generated X-band signal in the time domain; (b) the calculated instantaneous frequency-time diagram of the X-band signal; (c) a vertical-section of the instantaneous frequency-time diagram of the X-band signal; (d) waveform of the generated Ku-band signal in the time domain; (e) the calculated instantaneous frequency-time diagram of the Ku-band signal; (f) a vertical-section of the instantaneous frequency-time diagram of the Ku-band signal.

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Results of the X-band and Ku-band signals after digital autocorrelation are shown in Fig. 7. The insets give a zoom-in view of the two compressed signals. The width of both compressed pulses is 0.56 ns, corresponding to a pulse CR of 3.6×10⁵. The PSRs of the compressed X-band signal and the compressed Ku-band signal are 11.94 dB and 12.63 dB, respectively. The phase noise of the RF signal, flicker noise added by the active electrical amplifiers and low extinction ratio of the MZM restrict the PSR of the two compressed pluses. The discrepancy in the PSRs of the two compressed signals are mainly caused by clock asynchronization between the RF generator and the OEO. A photograph of the experimental setup is shown in Fig. 8.

Fig. 7. (a) Result of the X-band signal after digital autocorrelation; (b) Result of the Ku-band signal after digital autocorrelation; the insets show the peak of the compressed transmission signals.

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Fig. 8. Photograph of the experimental setup.

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Due to available electric mixers and antennas operating in X band and the similar characteristics of the multi-band signals, an experiment of detecting a moving target is presented to further evaluate the multi-band transmitter. Firstly, a radar system is built up. An electric mixer (HMC773ALC3B), a customized intermediate frequency (IF) amplifier with 25-dB power gain and an OSC make up an electric receiver. The developed multi-band radar transmitter, the electrical receiver and a pair of X-band horn antennas with power gain of 20 dB compose a radar system. Next, the test scenario is designed. The target is a trihedral corner reflector (TCR) attached to one end of a stick with a length of 2.5 m. The TCR and two antennas are placed on a horizontal plane. The TCR is making a pendulum-like motion on the same horizontal plane. The linear speed of the target is estimated by a simple test. The maximum value of the angular is measured as around π/4, and the temporal period of the moving target is measured as 4.6 seconds. Thus, the average linear speed of the objective can be calculated as around 1.7 m/s. The distance between the central position of the target’s trajectory and the radar transceiver is about 3 m, as shown in Fig. 9. Finally, a detection experiment is carried out.

Fig. 9. Schematic diagram of the experimental test scenario and photograph of the target.

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By adjusting the oscillation frequency as 12.4 GHz and setting frequency range of the input RF signal from 7 to 7.8 GHz, an LFMCW (frequency 9.2–10.8 GHz; temporal period 200 µs) waveform is generated from the transmitter, which is then split into two paths. The waveform along one path is directly emitted as a transmission signal; the waveform along other path, as a reference signal, is coupled into the receiver. The transmission signal is emitted by a transmitting antenna and reflected by the TCR. The echo signal is collected by a receiving atenna and subsequently de-chirped with the reference signal in a mixer, output of which is an IF signal. The IF signal is amplified by an IF amplifier and recorded by an OSC with a sampling rate of 20 MSa/s for further digital signal processing. The cable length difference is around 1.5 m between the reference signal and the echo signal.

When the IF signal is Fourier transformed in range direction, the variation of the target’s horizontal range is obtained, as shown in Fig. 10(a). The inset is a de-chirped result of the echo signal at a given moment. The PSR of the de-chirped signal is 11.81 dB. The range resolution of the radar is 0.08 m. Figure 10(b) shows the variation of the target’s horizontal velocity employing Fourier transform in velocity direction and short-time Fourier analysis. The velocity resolution of the radar is about 0.028 m/s. A simulation is also conducted and the result (in white curve) is basically in accord with the measured result, as shown in Fig. 10(b).

Fig. 10. (a) Measured TCR’s horizontal range variation with respect to time; inset shows the de-chirped result of the echo at a certain moment); (b) measured TCR’s horizontal velocity variation with respect to time (yellow curve); a simulation result of the TCR’s horizontal velocity variation with respect to time (white curve).

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4. Conclusion

To sum up, we present a novel reconfigurable microwave photonic multi-band radar transmitter with a wide frequency operating range. The radar transmitter is a solution to efficiently generating transmission signals with high quality and broad frequency tunable ranges. The developed radar transmitter works in six bands (from S to Ka) with a bandwidth of 1.6 GHz. Performances of the multi-band radar transmitter are evaluated via several proof-of-concept experiments. Range resolution of around 0.08 m and velocity resolution of around 0.028 m/s are obtained in the moving target detection experiment. The measured range and velocity information of the moving target are coincident with the actual results, and the experiment demonstrates the potential of the microwave photonic multi-band radar transmitter in practical radar applications.

Funding

National Natural Science Foundation of China (61690191, 61701476); National Key R&D Program of China (2018YFA0701900, 2018YFA0701901).

Disclosures

The authors declare no conflicts of interest.

References

1. M. Vespe, C. J. Baker, and H. D. Griffiths, “Automatic target recognition using multi-diversity radar,” IET Radar Sonar Navig. 1(6), 470–478 (2007). [CrossRef]

2. P. V. Dorp, R. Ebeling, and A. G. Huizing, “High-resolution radar imaging using coherent multiband processing techniques,” in Proceedings of Radar Conference (IEEE, 2010) pp. 981–986.

3. X. Wei, Y. Zheng, Z. Cui, and Q. Wang, “Multi-band SAR images fusion using the EM algorithm in Contourlet domain,” in Proceedings of International Conference on Fuzzy Systems & Knowledge Discovery (IEEE, 2007) pp. 502–506.

4. J. Capmany and D. Novak, “Microwave photonics combines two worlds,” Nat. Photonics 1(6), 319–330 (2007). [CrossRef]

5. J. Yao, “Microwave photonics,” in Proceedings of International Workshop on Electromagnetics; Applications and Student Innovation (iWEM) (IEEE2012), pp.314–335.

6. J. Chen, W. Zou, and W. Kan, “Reconfigurable microwave photonics radars,” in Proceedings of International Topical Meeting Microwave Photonics (IEEE, 2016), pp. 59–62.

7. P. Ghelfi, F. Laghezza, F. Scotti, G. Serafino, A. Capria, S. Pinna, D. Onori, C. Porzi, M. Scaffardi, A. Malacarne, V. Vercesi, E. Lazzeri, F. Berizzi, and A. Bogoni, “A fully photonics-based coherent radar system,” Nature 507(7492), 341–345 (2014). [CrossRef]

8. F. Scotti, D. Onori, and F. Laghezza, “Fully coherent S- and X-band photonics-aided radar system demonstration,” IEEE Microw. Wireless Compon. Lett. 25(11), 757–759 (2015). [CrossRef]

9. S. Melo, A. Bogoni, S. Pinna, F. Laghezza, and F. Scotti, “Dual-use system combining simultaneous active radar & communication, based on a single photonics-assisted transceiver,” in Proceedings of International Radar Symposium (IEEE, 2016), pp. 1–4.

10. Q. Guo, P. Zhou, and S. Pan, “Dual-band linear frequency modulation signal generation by optical frequency quadrupling and polarization multiplexing,” IEEE Photonics Technol. Lett. 29(16), 1320–1323 (2017). [CrossRef]

11. R. Li, W. Li, M. Ding, Z. Wen, Y. Li, L. Zhou, S. Yu, T. Xing, B. Gao, Y. Luan, Y. Zhu, P. Guo, Y. Tian, and X. liang, “Demonstration of a microwave photonic synthetic aperture radar based on photonic-assisted signal generation and stretch processing,” Opt. Express 25(13), 14334–14340 (2017). [CrossRef]

12. F. Zhang, Q. Guo, Z. Wang, P. Zhou, G. Zhang, J. Sun, and S. Pan, “Photonics-based broadband radar for high resolution and real-time inverse synthetic aperture imaging,” Opt. Express 25(14), 16274 (2017). [CrossRef]

13. J. Cao, R. Li, J. Yang, Z. Mo, J. Dong, X. Zhang, W. Jiang, and W. Li, “Photonic deramp receiver for dual-band LFM-CW radar,” J. Lightwave Technol. 37(10), 2403–2408 (2019). [CrossRef]

14. S. Peng, S. Li, X. Xue, X. Xiao, D. Wu, X. Zheng, and B. Zhou, “High-resolution W-band ISAR imaging system utilizing a logic-operation-based photonic digital-to-analog converter,” Opt. Express 26(2), 1978–1987 (2018). [CrossRef]

15. A. Wang, J. Wo, X. Luo, Y. Wang, W. Cong, P. Du, J. Zhang, B. Zhao, J. Zhang, Y. Zhu, J. Lan, and L. Yu, “Ka-band microwave photonic ultra-wideband imaging radar for capturing quantitative target information,” Opt. Express 26(16), 20708–20717 (2018). [CrossRef]

16. D. Eliyahu, D. Seidel, and L. Maleki, “Phase noise of a high performance OEO and an ultra low noise floor cross-correlation microwave photonic homodyne system,” in Proceedings of International Frequency Control Symposium (IEEE, 2008), pp. 811–814.

17. P. Ghelfi, D. Onori, F. Laghezza, and A. Bogoni, “Photonics for radars operating on multiple coherent bands,” J. Lightwave Technol. 34(2), 500–507 (2016). [CrossRef]

18. W. Li and J. Yao, “Optically tunable frequency-multiplying optoelectronic oscillator,” IEEE Photonics Technol. Lett. 24(10), 812–814 (2012). [CrossRef]

19. X. Liu, W. Pan, X. Hua, L. Yan, B. Luo, and B. Lu, “Investigation on tunable modulation index in the polarization-modulator-based optoelectronic oscillator,” IEEE J. Quantum Electron. 50(2), 68–73 (2014). [CrossRef]

20. X. S. Yao and L. Maleki, “Optoelectronic microwave oscillator,” J. Opt. Soc. Am. B 13(8), 1725–1735 (1996). [CrossRef]

21. X. S. Yao and L. Maleki, “Multi-loop optoelectronic oscillator,” IEEE J. Quantum Electron. 36(1), 79–84 (2000). [CrossRef]

22. X. S. Yao, L. Maleki, Y. Ji, G. Lutes, and M. Tu, “Dual-loop optoelectronic oscillator,” in Proceedings of International Frequency Control Symposium (IEEE, 1998), pp. 545–549.

23. W. Li and J. Yao, “A wideband frequency tunable optoelectronic oscillator incorporating a tunable microwave photonic filter based on phase-modulation to intensity-modulation conversion using a phase-shifted fiber Bragg grating,” IEEE Trans. Microw. Theory Techn. 60(6), 1735–1742 (2012). [CrossRef]

24. O. Lelièvre, V. Crozatier, P. Berger, G. baili, and G. Pillet, “A Model for Designing Ultralow Noise Single- and Dual-Loop 10-GHz Optoelectronic Oscillators,” J. Lightwave Technol. 35(20), 4366–4374 (2017). [CrossRef]

Reconfigurable multi-band microwave photonic radar transmitter with a wide operating frequency range

Abstract

1. Introduction

2. Principle of operation

2.1 Triple-loop OEO

2.2 Microwave photonic frequency-converting link

3. Experiments and results

4. Conclusion

Funding

Disclosures

References

Cited By

Figures (10)

Equations (5)

Optics Express