Open Access
Add to BibSonomy
Abstract
A leaky-wave antenna (LWA) based on reconfigurable spoof surface plasmon polaritons (SSPP) is proposed for beam scanning in the Ka band, which consists of a reconfigurable SSPP waveguide and a periodic array of metal rectangular split rings. Both numerical simulations and experimental measurements show that the reconfigurable SSPP-fed LWA has good performance in the frequency range from 25 to 30 GHz. Specifically, as the bias voltage changes from 0 to 15 V, we can achieve the maximum sweep range of 24° at a single frequency and 59° at multiple frequency points, respectively. Owing to the wide-angle beam-steering feature, as well as the field confinement and wavelength compression properties derived from the SSPP architecture, the proposed SSPP-fed LWA possesses great potential applications in the compact and miniaturized devices and systems of the Ka band.
© 2023 Optica Publishing Group under the terms of the Optica Open Access Publishing Agreement
1. Introduction
Leaky-wave antennas (LWAs), kinds of traveling-wave antennas, are widely used in radar and communication systems owing to the advantages of simple and compact structure, easy processing, easy integration, and etc. [1,2] Considerable LWAs based on different waveguide structures have been proposed yet, including microstrip lines, [3] substrate integrated waveguide (SIW), [4–7] composite right/left-handed transmission lines (CRLH-TL), [8,9] and SSPP. [10–15] Especially, reconfigurable LWAs have sparked a lot of interest. In 2018, Wang et al. proposed an electronically controlled beam-scanning LWA based on the architecture of corrugated microstrip line and theory of surface impedance modulation. [16] It is shown that the radiation direction of the leaky wave can be controlled in real time by changing the bias voltages of varactor diodes loaded onto the corrugated microstrip line. In 2019, a dual-band reconfigurable LWA was reported based on a corrugated microstrip line, in which, the surface impedance of both working modes can be adjusted through tuning the bias voltage of a varactor diode loaded to the unit cell of structure and the dual-band beam-scanning has been achieved. [17]
Recently, spoof surface plasmon polariton (SSPP) devices have been extensively investigated. [18–25] Among which, the SSPP-fed LWAs have been demonstrated with simple structures and excellent radiation performance, and furthermore, the unique dispersion characteristic of SSPP can provide new road towards the LWA design. [11,26] As a feed source, SSPP shows significant advantages. Firstly, due to the strong field confinements of SSPPs, the antennas will have a compact structure and suppressed mutual coupling with neighboring array elements or other devices, which indicates for the potential applications in highly integrated and miniaturized communication systems. Secondly, owing to the low loss characteristic, it can maintain good signal integrity and provide high gain for the antenna. [27] Thirdly, it is easy to obtain tunable and reconfigurable antennas by modulating the constituent units of the SSPP structure. Lastly, the groundless SSPP structure can also be used as a feeding source for antennas that need to reduce the influence of metallic ground. Despite the extensive studies of SSPP, reconfigurable SSPP-fed LWAs in the Ka band have not been reported yet. Additionally, the EM radiations in the Ka band show important applications in low-orbit satellite communications. It not only has the innate advantage of large bandwidth, but also can be perfectly integrated with various new technologies such as multipoint beam, high frequency multiplexing, and adaptive switching. [28,29] Thus, developing the SSPP LWAs in the Ka-band is of important significance, which can effectively combine the advantages of SSPP devices and Ka band.
In this work, we propose a reconfigurable LWA integrated with the SSPP waveguide as the feed. Specifically, the LWA consists of a reconfigurable SSPP waveguide and a periodic metal rectangular split-ring array, which are employed as the feeding source and the radiation structure of LWA, respectively. The dispersion characteristics of the SSPP unit can be adjusted by varying the capacitance of the varactor diodes loaded on the structure, thus allowing electronic control of the radiation angle and enabling beam scanning. Moreover, the radiations on both sides can be obtained due to the groundless structure. The radiation performance of the SSPP-fed LWA is investigated by both numerical simulations and experiments. Accordingly, the electrically controlled beam scanning property in the Ka band has been validated. We believe that the Ka-band SSPP-fed LWA possesses the potential applications in the minimized and highly integrated devices and systems due to the planar compact structure and good process compatibility.
2. Results
2.1 Design of the reconfigurable SSPP-fed LWA
The schematic structure of the proposed reconfigurable SSPP-fed LWA is shown in Fig. 1. As illustrated in Figs. 1(a) and 1(b), the LWA is composed of a reconfigurable SSPP waveguide and a split-ring antenna array located on the top surface of the dielectric substrate, and radio frequency (RF) high resistance lines placed on the bottom surface of the substrate. The substrate is Rogers RT5880 with a thickness of 0.254 mm, the relative permittivity of 2.2 and loss tangent of 0.0009. It is worth noting that the reconfigurable SSPP waveguide is composed of 21 reconfigurable SSPP units that are connected to two coplanar-waveguide feed ports via the gradient structure. As shown in Fig. 1(c) (the partially magnified image for SSPP unit and split-ring antenna element), each SSPP unit is loaded with a varactor diode, and the main geometric parameters are p = 2 mm, h = 1 mm, and w = 0.3 mm. The ring antenna array consists of eight metal split rings, and the center distance between adjacent rings is d = 5.1 mm, while the distance between the rings and the reconfigurable SSPP waveguide is optimized as g = 0.3 mm. We remark that the number of rings requires careful selection to achieve optimal energy radiation from a ring antenna array into free space. If the number of rings is too small, the ring array will fail to couple sufficient energy and the radiation intensity will be weak. Conversely, if the number of rings is too large, the rings away from the signal input port will be unable to couple the energy effectively, and the long transmission line may result in unnecessary loss. Hence, in this work, the ring antenna is optimized with eight rings. Since the bias circuits connected to the SSPP waveguide will distort the distributions of electric field and surface current, a RF isolator is designed to reduce this effect. In our scheme, a series of RF high-resistance lines composed of six 1.7 nH chip inductors and metallic strips are designed to shield the RF signal, which are connected to the reconfigurable SSPP unit via the through-hole (see Fig. 1(d)).
Fig. 1. Schematic diagram of the proposed reconfigurable SSPP-fed LWA. (a) The top layer structure and modulated radiation of the reconfigurable SSPP-fed LWA. (b) The bottom layer structure. (c) The partially magnified image of the SSPP units and the split-ring antenna elements with dimensions of p = 2 mm, h = 1 mm, d = 5.1 mm, w = 0.3 mm, and g = 0.3 mm. (d) Side view of the unit structure, in which, ${t_1}$= 0.035 mm, ${t_2}$= 0.254 mm.
2.2. Theory
According to some previous study, the reconfigurable SSPP unit structure can be equivalent to a circuit topology [30]. As illustrated in the inset of Fig. 2, the entire SSPP cell can be viewed as the combination of two symmetrical series branches (planar Goubau lines) and a shunt admittance (parallel-connected open-circuit coplanar waveguide). Then, a simple equation can be utilized to describe the dispersion relation based on the circuit topology and Bloch theorem
in which, ${k_m}$ and ${Z_m}$ are the equivalent wavenumber and impedance of the series branch, ${k_x}$ is the equivalent wavenumber of the whole structure, respectively, l is the equivalent length of the level branch, and p is the period length. Y is the admittance of the shunt branch, which can be obtained byFig. 2. (a) The dispersion curves of the reconfigurable SSPP unit loaded with different DC bias voltages. The inset presents the equivalent circuit topology of the SSPP unit. (b) Schematic diagram of the reconfigurable SSPP-fed LWA for beam scanning, where the geometric parameters of the split-ring are ${w_1}$= 3.2 mm, ${w_2}$= 2.6 mm, and ${w_3}$= 0.4 mm.
Based on the aforementioned theoretical analysis, it is clear that the wavenumber ${k_x}$ of the SSPP waveguide structure can be controlled by the bias voltage, which is essential for achieving the reconfigurable SSPPs and the beam scanning. Furthermore, we obtained the dispersion curves of the reconfigurable SSPP unit cell by loading different DC bias voltages onto the varactor diode. [31] It is observed from Fig. 2(a) that the dispersion curves of the SSPP unit deviate from the light line (dispersion relation of waves in vacuum), showing the characteristic of slow wave. Additionally, the asymptotic frequency of the SSPP unit exhibits a blue shift from 31.7 to 35.4 GHz as the bias voltage grows from 0 to 15 V. Then, if the operating frequency is fixed, the dispersion relation of the reconfigurable SSPP waveguide will be electrically modulated by using an external bias voltage.
Next, the split-ring arrays are placed around the SSPP waveguide, and then, the EM wave will be easily coupled to the ring array, which excites the ring antenna array and then radiates the electromagnetic energy into free space. The radiation schematic is displayed in Fig. 2(b). Obviously, when the EM wave propagates along the SSPP waveguide, the phase differs at different locations on the waveguide, and the phase difference between adjacent rings can be deduced from $\varDelta \psi = {k_x}d$. While the phase difference of EM radiations from the adjacent metal split rings to the tilted equal-phase surface is noted as $\varDelta \varphi = {k_0}\Delta \textrm{s = } - {k_0}d\sin \theta $, where ${k_0}$ and $\Delta s$ denote the wavenumber and optical path difference in free space, respectively, and d is the distance between the adjacent rings. Considering that, for the fundamental mode of SSPP waveguide, kx > k0, then, the phase difference and beam direction θ should satisfy the following relationship: [11]
Then, substitute Eqs. (1) and (2) into (3), we can get:
This equation approximately describes the relationship among the deflection angle θ of the radiated beam, the shunt admittance (which is the function of the working frequency ω and the tunable capacitance C), and the ring period d. And hence, it is an approximate but very intuitive relation to understand the working principle of the beam scanning antenna. That is, as the capacitance of the varactor diode is tuned through applying a bias voltage, the wavenumber ${k_x}$ of the SSPP wave will change along with it, and thus the radiation direction of the antenna will follow this change to keep the phase relation. It is worth noting that, to simplify the analysis, the calculation method here ignores the parasitic resistance of the varactor diode, the mismatch caused by the feed network, the difference in the directional diagram due to the edge effect and the mutual coupling of the antenna elements.
2.3 Results and discussion
Subsequently, the performances of the SSPP-fed LWA were investigated both numerically and experimentally. Full-wave simulations of the LWA were performed in CST Microwave Studio. The S-parameters of the LWA were measured using a vector network analyzer (Agilent N5230C). Figures 3(a)–3(b) present the photo of the fabricated sample of the SSPP-fed LWA and the measurement setup. The varactor type is MA46H120. As listed in Table 1, when the reverse bias voltage changes from 0 to 15 V, the capacitance can be continuously adjusted from 1.226 to 0.178 pF.
Fig. 3. (a) Photos of the reconfigurable SSPP-fed LWA. (b) Experimental setup of the SSPP-fed LWA in a microwave anechoic chamber.
Table 1. Capacitance values of varactor diode at different voltages
Figures 4(a)–4(b) show the simulated and measured results of the S-parameters between 20 and 40 GHz. We notice from the simulation curves that there is a remarkable blue shift of the cutoff frequency of SSPP waveguide with the increasing bias voltage. Similar tendency can be observed from the measured curves, although the cutoff characteristic gets not so sharp. We also observe that the measured cutoff frequency is smaller than the corresponding simulation value, which can be possibly ascribed to the fabrication errors, parasitic capacitance induced by the packaging and welding processes of the varactor diodes, and the deviation of dielectric parameters of the substrate. Moreover, there is a strong reflection peak before the cutoff frequency for the measured results, which is possibly caused by the so-called OSB (open-stopband) effect. At 0 V, the reflection peak appears around 28.5 GHz. Then, as the voltage increases to 7 V and 15 V, the peak position undergoes a blue shift, that is, OSB effect works in the higher frequency region. Nevertheless, the reflection peak is not observed in the simulated S11 curve since the parasitic effect of components cannot be accurately quantified. The influence of the OSB effect will be discussed in detail later. Except for the reflection peak induced by the OSB effect, the reflection coefficients S11 from 25 to 30 GHz is all less than −10 dB, which indicates a good impedance and momentum matching. It is also observed that the simulated and measured results of S21 under different capacitances of varactor diodes are all less than −10 dB over the radiation frequency band (25-30 GHz), which indicates a high radiation efficiency. Moreover, the measured S21 is lower than the simulated one due to the increased reflection coefficient caused by the OSB effect, unpredictable losses in the bias circuit, and etc. [32,33] In order to further analyze the radiation process, we simulated the electric field distribution at 0 V and 15 V at 30 GHz, as shown in Figs. 4(c)–4(d). As the transmission energy of the reconfigurable SSPP waveguide gradually decreases, a backward-scanning beam is formed.
Fig. 4. (a-b) The simulated and experimental S-parameters of LWA in the frequency range of 20-40 GHz under different bias voltages. (c-d) The simulated electric field distribution in the xoz plane at 30 GHz with bias voltages of 0 V and 15 V respectively.
The far-field radiation patterns of LWA were characterized using an antenna measurement system in the microwave anechoic chamber, which consists of a signal source (KEYSIGHT E8257D), a standard gain horn antenna with the bandwidth of 25-40 GHz and gain of 25 dBi, and a spectrum analyzer (KEYSIGHT N9040B). A 50-Ω load was loaded on the right end of the SSPP waveguide for the impedance matching. Figure 5 illustrates the simulated and measured normalized far-field radiation patterns of the reconfigurable SSPP-fed LWA between 25 and 30 GHz. The corresponding beam radiation angles are summarized in Table 2. The diagonal lines indicate some radiation angles that cannot be measured accurately. Due to the groundless structure, the antenna radiates on both sides. At 25 GHz, when the voltage increases from 0 to 15 V, the simulated radiation angle scans from -45° to -54°, and the tested radiation angle steers from -42° to -57° (Figs. 5(a)–5(b)). The simulated and tested radiation patterns are generally in good agreement, nevertheless the tested beam scanning range is slightly larger than the simulated result. As the frequency increases to 26 GHz (Figs. 5(c)–5(d)), the simulated and measured radiation angle changes from -35° to -48°, and from -27° to -54°, respectively. The increased steering range is due to the fact that the variation value of the wavenumber of the reconfigurable SPP unit increases with frequency, and according to Eq. (4), the scanning range will also increase. At 27 GHz shown in Figs. 5(e)–5(f), the measured radiation intensity at 0 V is greatly reduced due to the ohmic loss, [16,31] and no obvious main beam can be observed. The ohmic loss is possibly caused by the parasitic resistance of the varactor diode, which will increase as the bias voltage decreases, consequently, ohmic loss will be more significant in the low voltage region especially at 0 V. As shown in Figs. 6(a)–6(f), the maximum scanning range of 24° can be obtained at 30 GHz. However, in addition to the ohmic loss, the OSB effect also affects the radiation efficiency when the beam is scanned to the broadside. Therefore, we cannot accurately measure the radiation angle in the low voltage case since the main beam cannot be clearly distinguished with side lobes. It is in coincidence with the measured S-parameter results in Fig. 4(b) that when the voltage is 0 V, S11 parameter has a high reflection between 28 and 30 GHz, which will lead to the remarkable decrease of the antenna radiation at these frequencies. Herein, we can conclude that a simulated sweep range of 24° at 30 GHz and 59° at multiple frequency points can be attained. The experimental single-frequency beam steering range can reach up to 27° at both 26 and 30 GHz despite the influence of the OSB effect and the ohmic loss, while the corresponding value at multiple frequency points is 57°.
Fig. 5. The simulated and measured results of the far-field normalized radiation patterns between 25 and 27 GHz.
Fig. 6. The simulated and measured results of the far-field normalized radiation patterns between 28 and 30 GHz.
Table 2. Simulated and measured angles of the radiation beams
Next, the factors influence the radiation performances will be illustrated more quantitatively. OSB is a kind of impact that antenna cannot effectively realize the side emission, i.e., part of the incident wave reflects, part of the wave exponentially decays, and hence the gain drops sharply, and its beam scanning area is cut into two discontinuous intervals [32,34,35]. Here, the influence of the inductance on the OSB effect are discussed in detail. Figure 7(a) illustrates that when the regulation voltage is 7 V and the inductance of the RF high resistance line series is 10 nH, a strong reflection peak can be observed near 32 GHz (see the red solid line in Fig. 7(a)). As the inductance decreases, the strong reflection peak generated by the OSB phenomenon shows a blue shift and the reflection coefficient gradually rises, and when the working frequency is getting close to the reflection peak, the radiation gain is greatly reduced. Nevertheless, the reflection peaks appear below 30 GHz in experiment (see Fig. 4(b)), and thus the OSB effect works over 28-30 GHz, although it is designed beyond this frequency region.
Fig. 7. (a) The effect of series inductance of RF high resistance line on S-parameters when the voltage is 7 V. (b) The radiation gain versus voltage under different parasitic resistance values.
Additionally, the parasitic resistance brought by the varactor diode will also play a role in the reduction of radiation efficiency. [36] We simulated the effect of different parasitic resistances on the radiation gain of the reconfigurable SSPP-fed LWA. As shown in Fig. 7(b), with the increase of parasitic resistance, the radiation gain shows a significant decrease and the effect is more pronounced at lower voltages. Specifically, the average radiation gain decreases from 10.3 to 6.45 dBi as the parasitic resistance increases from 0 to 10 Ω at 0 V. Besides, as the operation frequency increases, the influence of the parasitic resistance on the radiation gain becomes more and more remarkable, which can be probably attributed to the enhanced field confinement and increased sensitivity to ohmic loss at high frequencies. Fortunately, the parasitic resistance hardly influences the radiation angle. Thus, we may draw a conclusion that the measured low radiation intensity of the LWA in the low-voltage portion can be ascribed to the ohmic loss induced by the parasitic resistance of the varactor diode, and that for the broadside radiation case, the gain reduction is owing to the joint effect of the OSB phenomenon and the ohmic loss. To improve the radiation efficiency and gain of the SSPP-fed LWA, we can use varactor diodes with lower parasitic resistance, and also employ the impedance transformation method [32] or introduce asymmetric structure [36,37] to suppress the OSB phenomenon.
3. Conclusion
In summary, we proposed a Ka-band LWA using a reconfigurable SSPP waveguide to provide feed. The SSPP waveguide is composed of periodic SSPP unit cells loaded with varactor diodes, and hence, the wave vector of the SSPP waveguide is tunable, resulting in a reconfigurable radiation pattern of the LWA. Both the simulated and measured results show that the reconfigurable SSPP-fed LWA has good beam scanning performance in the frequency range from 25 to 30 GHz. Specifically, as the bias voltage changes from 0 to 15 V, a maximum sweep range of 24° at a single frequency point and 59° at multiple frequency points can be achieved. The antenna developed in this work has a simple design, low cost, satisfied beam sweeping performance, and good process compatibility, which has potential applications in highly integrated and miniaturized communication systems such as radar and satellite communication systems.
Funding
National Key Research and Development Program of China (2017YFA0700201, 2017YFA0700202, 2017YFA0700203, 2018YFB1801505); Southeast University “Xiaomi Young Scholars” Program; National Natural Science Foundation of China (62071117, 62288101); 111 Project (111-2-05); State Key Laboratory of Millimeter Waves (K202316).
Acknowledgments
This work was supported by the National Key Research and Development Program of China (2018YFB1801505, 2017YFA0700201, 2017YFA0700202, and 2017YFA0700203), Southeast University “Xiaomi Young Scholars” Program, National Natural Science Foundation of China (62288101 and 62071117), 111 Project (111-2-05), and the State Key Laboratory of Millimeter Waves (K202316).
Authors’ Contributions. Qi Chen: Methodology (lead); Validation (lead); Writing-original draft preparation (lead). Xiaojian Fu: Conceptualization (lead); Supervision (lead); Writing-review and editing (lead). Jiang Luo: Resources (lead). Yuan Fu: Validation (equal). Yujie Liu: Validation (supporting). Lei Shi: Validation (supporting). Fei Yang: Validation (supporting). Hao Chi Zhang: Writing-review and editing (supporting). Hui Feng Ma: Writing-review and editing (supporting). Tie Jun Cui: Supervision (equal); Writing-review and editing (lead).
Disclosures
The authors declare no conflicts of interest.
Data availability
Data that support the findings of this study are available from the corresponding authors upon reasonable request.
References
1. S. Gupta, S. Abielmona, and C. Caloz, “Microwave Analog Real-Time Spectrum Analyzer (RTSA) Based on the Spectral–Spatial Decomposition Property of Leaky-Wave Structures,” IEEE Trans. Microwave Theory Tech. 57(12), 2989–2999 (2009). [CrossRef]
2. F. M. Monavar, S. Shamsinejad, R. Mirzavand, J. Melzer, and P. Mousavi, “Beam-Steering SIW Leaky-Wave Subarray With Flat-Topped Footprint for 5 G Applications,” IEEE Trans. Antennas Propag. 65(3), 1108–1120 (2017). [CrossRef]
3. W. Hong, T.-L. Chen, C.-Y. Chang, J.-W. Sheen, and Y.-D. Lin, “Broadband tapered microstrip leaky-wave antenna,” IEEE Trans. Antennas Propag. 51(8), 1922–1928 (2003). [CrossRef]
4. Y. J. Cheng, W. Hong, and K. Wu, “Millimetre-wave monopulse antenna incorporating substrate integrated waveguide phase shifter,” IET Microw. Antennas Propag 2(1), 48–52 (2008). [CrossRef]
5. F. Xu, K. Wu, and X. Zhang, “Periodic Leaky-Wave Antenna for Millimeter Wave Applications Based on Substrate Integrated Waveguide,” IEEE Trans. Antennas Propag. 58(2), 340–347 (2010). [CrossRef]
6. J. Liu, D. R. Jackson, and Y. Long, “Substrate Integrated Waveguide (SIW) Leaky-Wave Antenna With Transverse Slots,” IEEE Trans. Antennas Propag. 60(1), 20–29 (2012). [CrossRef]
7. E. Torabi, D. Erricolo, P.-Y. Chen, W. Fuscaldo, and R. Beccherelli, “Reconfigurable beam-steerable leaky-wave antenna loaded with metamaterial apertures using liquid crystal-based delay lines,” Opt. Express 30(16), 28966–28983 (2022). [CrossRef]
8. C. Caloz and T. Itoh, “Array factor approach of leaky-wave antennas and application to 1-D/2-D composite right/left-handed (CRLH) structures,” IEEE Microw. Wireless Compon. Lett. 14(6), 274–276 (2004). [CrossRef]
9. F. P. Casares-Miranda, C. Camacho-Penalosa, and C. Caloz, “High-gain active composite right/left-handed leaky-wave antenna,” IEEE Trans. Antennas Propag. 54(8), 2292–2300 (2006). [CrossRef]
10. J. J. Xu, J. Y. Yin, H. C. Zhang, and T. J. Cui, “Compact Feeding Network for Array Radiations of Spoof Surface Plasmon Polaritons,” Sci. Rep. 6(1), 22692 (2016). [CrossRef]
11. J. Y. Yin, J. Ren, Q. Zhang, H. C. Zhang, Y. Q. Liu, Y. B. Li, X. Wan, and T. J. Cui, “Frequency-Controlled Broad-Angle Beam Scanning of Patch Array Fed by Spoof Surface Plasmon Polaritons,” IEEE Trans. Antennas Propag. 64(12), 5181–5189 (2016). [CrossRef]
12. Q. Zhang, Q. Zhang, and Y. Chen, “Spoof Surface Plasmon Polariton Leaky-Wave Antennas Using Periodically Loaded Patches Above PEC and AMC Ground Planes,” Antennas Wirel. Propag. Lett. 16, 3014–3017 (2017). [CrossRef]
13. D.-F. Guan, P. You, Q. Zhang, Z.-H. Lu, S.-W. Yong, and K. Xiao, “A Wide-Angle and Circularly Polarized Beam-Scanning Antenna Based on Microstrip Spoof Surface Plasmon Polariton Transmission Line,” Antennas Wirel. Propag. Lett. 16, 2538–2541 (2017). [CrossRef]
14. Q. Zhang, Q. Zhang, and Y. Chen, “High-efficiency circularly polarised leaky-wave antenna fed by spoof surface plasmon polaritons,” IET Microw. Antennas Propag. 12(10), 1639–1644 (2018). [CrossRef]
15. P. Ge, Z. Wang, X. Tu, M. Lu, G. Fan, and B. Xiao, “A dual-band frequency scanning antenna based on spoof SPPs transmission line,” Opt. Commun. 524, 128743 (2022). [CrossRef]
16. M. Wang, H. F. Ma, H. C. Zhang, W. X. Tang, X. R. Zhang, and T. J. Cui, “Frequency-Fixed Beam-Scanning Leaky-Wave Antenna Using Electronically Controllable Corrugated Microstrip Line,” IEEE Trans. Antennas Propag. 66(9), 4449–4457 (2018). [CrossRef]
17. M. Wang, H. F. Ma, W. xuan Tang, H. C. Zhang, W. xiang Jiang, and T. J. Cui, “A Dual-Band Electronic-Scanning Leaky-Wave Antenna Based on a Corrugated Microstrip Line,” IEEE Trans. Antennas Propag. 67(5), 3433–3438 (2019). [CrossRef]
18. S. Sun, Q. He, S. Xiao, Q. Xu, X. Li, and L. Zhou, “Gradient-index meta-surfaces as a bridge linking propagating waves and surface waves,” Nat. Mater. 11(5), 426–431 (2012). [CrossRef]
19. X. Shen, T. J. Cui, D. Martin-Cano, and F. J. Garcia-Vidal, “Conformal surface plasmons propagating on ultrathin and flexible films,” Proc. Natl. Acad. Sci. U.S.A . 110(1), 40–45 (2013). [CrossRef]
20. H. F. Ma, X. Shen, Q. Cheng, W. X. Jiang, and T. J. Cui, “Broadband and high-efficiency conversion from guided waves to spoof surface plasmon polaritons,” Laser Photonics Rev. 8(1), 146–151 (2014). [CrossRef]
21. A. Kianinejad, Z. N. Chen, and C.-W. Qiu, “A Single-Layered Spoof-Plasmon-Mode Leaky Wave Antenna With Consistent Gain,” IEEE Trans. Antennas Propag. 65(2), 681–687 (2017). [CrossRef]
22. X. Zhang, W. Y. Cui, Y. Lei, X. Zheng, J. Zhang, and T. J. Cui, “Spoof Localized Surface Plasmons for Sensing Applications,” Adv. Mater. Technol. 6(4), 2000863 (2021). [CrossRef]
23. Y. Ren, J. Zhang, X. Gao, X. Zheng, X. Liu, and T. J. Cui, “Active spoof plasmonics: from design to applications,” J. Phys.: Condens. Matter 34(5), 053002 (2022). [CrossRef]
24. Z. W. Cheng, M. Wang, Z. H. You, H. F. Ma, and T. J. Cui, “Spoof surface plasmonics: principle, design, and applications,” J. Phys.: Condens. Matter 34(26), 263002 (2022). [CrossRef]
25. H. Yan, L. Jing, J. Zhao, C. Niu, Y. Zhang, L. Du, and Z. Wang, “Broadband nonreciprocal spoof plasmonic phase shifter based on transverse Faraday effects,” Opt. Express 30(13), 24000–24008 (2022). [CrossRef]
26. D. Liao, Y. Zhang, and H. Wang, “Wide-Angle Frequency-Controlled Beam-Scanning Antenna Fed by Standing Wave Based on the Cutoff Characteristics of Spoof Surface Plasmon Polaritons,” Antennas Wirel. Propag. Lett. 17(7), 1238–1241 (2018). [CrossRef]
27. H. C. Zhang, Q. Zhang, J. F. Liu, W. Tang, Y. Fan, and T. J. Cui, “Smaller-loss planar SPP transmission line than conventional microstrip in microwave frequencies,” Sci. Rep. 6(1), 23396 (2016). [CrossRef]
28. I. del Portillo, B. G. Cameron, and E. F. Crawley, “A technical comparison of three low earth orbit satellite constellation systems to provide global broadband,” Acta Astronaut. 159, 123–135 (2019). [CrossRef]
29. P. Capodieci, F. Martinino, M. Melani, O. Michelangeli, E. Salvatori, S. Cacopardi, and G. Reali, “Use of 20/30 GHz frequencies for broadband satellite communication systems,” in N. Ohta, ed. (1993), pp. 14–30.
30. H. C. Zhang, T. J. Cui, J. Xu, W. Tang, and J. F. Liu, “Real-Time Controls of Designer Surface Plasmon Polaritons Using Programmable Plasmonic Metamaterial,” Adv. Mater. Technol. 2(1), 1600202 (2017). [CrossRef]
31. X. Zhang, H. C. Zhang, W. X. Tang, J. F. Liu, Z. Fang, J. W. Wu, and T. J. Cui, “Loss Analysis and Engineering of Spoof Surface Plasmons Based on Circuit Topology,” Antennas Wirel. Propag. Lett. 16, 3204–3207 (2017). [CrossRef]
32. S. Paulotto, P. Baccarelli, F. Frezza, and D. R. Jackson, “A Novel Technique for Open-Stopband Suppression in 1-D Periodic Printed Leaky-Wave Antennas,” IEEE Trans. Antennas Propag. 57(7), 1894–1906 (2009). [CrossRef]
33. P. Burghignoli, G. Lovat, and D. R. Jackson, “Analysis and Optimization of Leaky-Wave Radiation at Broadside From a Class of 1-D Periodic Structures,” IEEE Trans. Antennas Propag. 54(9), 2593–2604 (2006). [CrossRef]
34. S. Otto, A. Al-Bassam, A. Rennings, K. Solbach, and C. Caloz, “Radiation Efficiency of Longitudinally Symmetric and Asymmetric Periodic Leaky-Wave Antennas,” Antennas Wirel. Propag. Lett. 11, 612–615 (2012). [CrossRef]
35. Y. Li, Q. Xue, E. K.-N. Yung, and Y. Long, “The Periodic Half-Width Microstrip Leaky-Wave Antenna With a Backward to Forward Scanning Capability,” IEEE Trans. Antennas Propag. 58(3), 963–966 (2010). [CrossRef]
36. S. Otto, A. Al-Bassam, A. Rennings, K. Solbach, and C. Caloz, “Transversal Asymmetry in Periodic Leaky-Wave Antennas for Bloch Impedance and Radiation Efficiency Equalization Through Broadside,” IEEE Trans. Antennas Propag. 62(10), 5037–5054 (2014). [CrossRef]
37. S. Otto, Z. Chen, A. Al-Bassam, A. Rennings, K. Solbach, and C. Caloz, “Circular Polarization of Periodic Leaky-Wave Antennas With Axial Asymmetry: Theoretical Proof and Experimental Demonstration,” IEEE Trans. Antennas Propag. 62(4), 1817–1829 (2014). [CrossRef]
