1. Introduction

Wireless communication and polarization conversion of EM waves are crucial in engineering applications. Polarization conversion metasurfaces (PCMs) are functional devices that manipulate the direction of incident EM waves [1,2]. While engineered PCMs excel in radar cross-section (RCS) reduction and gain enhancement, they may not optimize EM signal reception or transmission efficiency. Polarization control is a critical aspect of EM waves and finds applications in various fields, including wireless communications [3], antenna design [4], fiber optics, and sensing systems [5]. Numerous polarization conversion metasurfaces (PCMs) have been developed, categorized based on their specific functionalities, such as reflecting type [6], transmitting type [7], cross-polarization conversion [8], linear-to-circular conversion [9], and circular-to-linear conversion [9]. However, most reported PCMs are limited to a single functionality, meaning that the designed structures can only control and alter the polarization of incident waves.

Recently, a few researchers have designed multi-functional structures that are capable of performing multiple functions [10–15]. For instance, in [11], a multi-functional polarization conversion metasurface (PCM) was developed to efficiently convert incident linearly polarized EM waves to their orthogonal counterparts and circularly polarized waves across two frequency bands. Similarly, in [13,14], a multi-functional metasurface was designed with the ability to switch between absorption and polarization conversion. Furthermore, in [15,16], multi-functional polarization transforming metasurfaces were created to transform linearly polarized waves into either linear or circular polarized reflected waves.

To the best of the author’s knowledge, the design and investigation of a multi-functional structure incorporating controllable polarization conversion and EM radiation have not been explored extensively. Most research in this area focuses on designing polarization conversion metasurfaces (PCMs) and integrating them with antennas to reduce the radar cross-section (RCS) [17,18]. In [18], a dual circularly polarized (CP) Fabry-Perot (FP) resonator antenna was developed, utilizing a PCM as the superstrate for the FP resonator antenna. Similarly, in [17], PCMs were designed and loaded around the antenna to achieve a low RCS.

However, there have been a few attempts to design absorbing metasurfaces that exhibit both EM absorption and radiation characteristics [19–24]. These absorbing structures typically incorporate resistive lumped components to enhance absorption, but this can adversely affect radiation performance. Moreover, these designed absorbing structures have some notable shortcomings, including the lack of a polarization conversion function, limited RCS reduction bandwidth, and low gain.

In this study, we propose the design of a multi-functional metasurface capable of efficient polarization conversion and EM radiation. The metasurface has the ability to convert incident x- or y-polarized waves into cross-polarized reflected waves within the frequency range of 8.0 to 13.5 GHz. Additionally, the metasurface functions as a metasurface antenna array, exhibiting high gain and low radar cross-section (RCS). It operates at two resonance frequency bands, specifically 8.6 GHz and 12.4 GHz, with a maximum peak gain of 15.1 dBi and 16.4 dBi at 8.6 GHz and 12.4 GHz, respectively. To achieve wideband RCS reduction, two different methods have been employed, addressing both the propagation direction and other directions.

2. Design and analysis of the structure

To obtain the capability of performing multiple functions by a single surface, we need to design different layers of the proposed structure carefully such that they can exhibit distinguished behaviors simultaneously. The surface topology can be made to behave as a polarization converter or a radiator by selecting the input source. The multiple modes of the proposed metasurface are displayed in Fig. 1(a). It can be observed from the Fig. 1(a), that the metasurface can convert an incident x-polarized wave into a y-polarized reflected wave. The same surface also functions as a radiator when all elements on the top layer are excited by some source signal. Moreover, the same surface also acts as a low scattering platform for incoming energy within the wideband.

 

Fig. 1. (a) Operational modes of the structure, (b) the geometry of the unit cell, (c) co-polarization, cross-polarization reflection coefficients and polarization conversion ratio (PCR) of the proposed structure.

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3. Polarization conversion mode

3.1 PCM unit cell design

The schematic view of the proposed unit cell is displayed in Fig. 1(b). The proposed unit cell is composed of multiple layers has two F4B substrates. A circular patch within a split ring is printed on the fully grounded substrate having a thickness of $h_1=$ $3mm$ with the dielectric constant being $2.65$. The circular ring is split diagonally to ensure the polarization-insensitive behavior of the unit cell. The unit cell is simulated with the computer simulation technology (CST) Microwave Studio 2019. In the simulation, along x- and y-directions the periodic boundaries were considered and the unit cell was excited by Floquet port to evaluate its frequency-dependent response in the infinite periodic structure. The optimized dimensions of the proposed unit cells are as follows. p=$13$mm, wp=$0.61$ mm, wc = $0.72$mm Rp1= $4.4$ mm Rp2 = $2.7$mm, h2 = $1$mm, and t = $0.035$mm.

3.2 Mechanism of polarization conversion

From any reflective EM surface, the reflected EM wave is composed of two field components. One of them has the same polarization as the incident wave which is called a co-polarized reflected wave. The other component which is orthogonal to the incident wave is called cross-polarized reflected wave [1]. To analyze the co- and cross-polarization, we are using the Jones reflection matrix $R$ given by.

The reflection coefficients of the co-polarized and cross-polarized reflected waves are shown in Fig. 1(c) for the case in which x-polarized field $E_{i}= \hat {x}E_{\circ }e^{ikz}$ is normally incident i.e., at $0^{\circ }$ on the metasurface. It shows that the designed surface has a wideband conversion frequency range i.e., from $8.0$ GHz to $13.5$GHz. The magnitude of the co-polarized reflected wave is less than -$10$ dB and that of the cross-polarized reflected wave is nearly equal to $0$dB. A larger cross-polarized reflected wave than a co-polarized reflected wave ensures that x-polarized incident wave is completely converted into y-polarized reflected wave within the frequency range from 8GHz to $13.5$GHz.

Polarization conversion ratio (PCR) is a good measure of cross-polarization conversion that can be defined for an incident x-polarization as:

According Eq. (2), the PCR shown in Fig. 1(c), is plotted against resonance frequencies. As can be seen that the proposed design gives an high PCR within resonance frequencies. In Fig. 1(c), the PCR is more than $97{\% }$ in the frequency range from $8.0$ GHz to $13.5$ GHz, indicating that the proposed structure functions as an efficient linear polarizing metasurface. Furthermore, the PCR attains its maximum value approaching $100{\% }$ at resonance frequencies of $8.35$ GHz, $10.55$ GHz, and $12.70$ GHz.

3.3 Theoretical analysis

To better understand the phenomenon of polarization conversion, we have to find the eigenvalues and eigen-polarizations of the proposed structure. To carry out this theoretical analysis for the sake of convenience let us ignore the dielectric losses. For an efficient polarization conversion metasurface, the magnitude of co-polarized reflection coefficient is nearly equal to zero i.e., $|R_{yx}|=|R_{xx}|\approx 0$ and cross-polarized reflection coefficient is nearly equal to one i.e., $|R_{yx}|=|R_{xy}|\approx 1$. Let’s substitute these values in reflection coefficient matrix (R) given in Eq. (1)

The independent eigenvector for the matrix given in Eq. (3) are $u = \begin {pmatrix} 1 & 1 \end {pmatrix}^{T}$, and $v = \begin {pmatrix} -1 & 1 \end {pmatrix}^{T}$. Corresponding to the eigenvector $u = \begin {pmatrix} 1 & 1 \end {pmatrix}^{T}$ the eigenvalue is $e^{jo}=1$ which means that an incident u-polarized wave is reflected with unity magnitude and $0^{\circ }$ phase. Similarly, for the eigenvector $v = \begin {pmatrix} -1 & 1 \end {pmatrix}^{T}$ the eigenvalue of $e^{jo}=-1$ which indicates that an incident v-polarized wave reflected with unity magnitude and $180^{\circ }$ phase. For both u- and v- polarized incident waves there is no cross-polarization conversion. Therefore, we have $|R_{uu}|=|R_{vv}| \approx 1$ and $|R_{uv}|=|R_{vu}| \approx 0$. Let consider a y-polarized EM wave with E-field $E_{i}=\hat {y}E_{i}e^{jkz}$ striking the surface as depicted in Fig. 2(a). In Fig. 2(a), there are two coordinate systems i.e., xy coordinate system and uv coordinate system, the u- axis is $+45^{\circ }$ anti-clock wise rotated to the x-axis and similarly, v-axis is rotated at $-45^{\circ }$ to y-axis. It can be found from Fig. 2(a), the unit cell geometry is anisotropic along u- and v-axis. Also, the unit cell is symmetric along u- and v-axis. Now let’s resolve the incident E-field along y-axis into two orthogonal components i.e., $E_{i}=\hat {y}E_{i} =\hat {u}E_{iu}+\hat {v}E_{iv}$ at position $z=0$ where $E_{iu}=0.707E_{i}$ and also $E_{iv}=0.707E_{i}$. As the $u$- and $v$-polarized incident waves reflected with same magnitude and $0^{\circ }$ and $180^{\circ }$ in phase, therefore the total reflected E-field will become;

From Eq. (4), the reflected field is along x-axis ensuring that cross polarization occurs. From Fig. 2(a), the $E_{r}$ is shown be along the x-axis that is generated from the addition of $E_{ru}$ and $E_{ru}$.

 

Fig. 2. (a) Decomposition of E-field along y-axis into $u$ and $v$ components and magnitude of co-polarized reflection coefficients and (b) their phases and (c) the simulated surface current distribution of proposed unit.

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In above discussion, the phases of the reflected u- and v-components the phases were $0^{\circ }$ and $180^{\circ }$ respectively. However, in general, the phase difference should be $180^{\circ }$, i.e., $\phi _{uu}-\phi _{vv} =180^{\circ }$. Which means that for cross-polarization conversion the requirement is the orthogonal components of the incident wave should reflect with magnitude equal to 1 and the phase difference equal to $180^{\circ }$. To verify this, we consider a y-polarized incident E-field, $E_{i}=\hat {y}E_{iu}=\hat {u}E_{iu}+\hat {v}E_{iv}$ at $z=0$, $E_{iu}=0.707E_{i}$ Then the total reflected field will become:

As we know the co-polarization component has unit magnitude i.e., $|R_{uu}|=|R_{vv}|=1$ and also $|R_{uv}|=|R_{vu}|=0$ . Therefore Eq. (5) becomes:

If the incident and reflected E-fields are orthogonal then $E_i.E_r=0$, which gives:

The solution of Eq. (7) is non-trivial if $\varphi _{uu}-\varphi _{uu}= \pi$, hence it is proved that the reflected wave has a phase difference equal to 180-degree. To verify the above theoretical analysis, we have simulated the designed unit for an incident u- and v-polarized waves as shown in Fig. 2(a). One can find that the magnitude of the co-polarized reflected wave is close to 1 i.e., $|R_{uu}|=|R_{vv}|>0.99$ within the frequency band of $8.0$ GHz to $13.5$ GHz. The phases of the co-polarized reflected waves are also shown in Fig. 2(b), the phase difference is nearlly 180-degree.

3.4 Surface current distribution

To verify the mechanism of polarization conversion in the designed unit cell, we simulated the current distributions on the top and ground layers at two resonance frequencies: 8.5 GHz and 12 GHz as shown in Fig. 2(c). At 8 GHz, the currents on the metallic split ring flow in opposite directions compared to the surface current distribution on the grounded bottom layer, indicating a magnetic dipole resonance. This is supported by the increase in permeability ($\mu$), resulting in a higher surface impedance ($\eta = \sqrt {{{\mu }/\varepsilon }}$) and in-phase reflection. Additionally, the currents on the edge of the circular patch flow in the same direction on the ground, signifying an electric dipole resonance. Similarly, at 12 GHz, the currents on the split ring flow in the same direction as the current on the ground, representing an electric dipole-dipole resonance. The magnetic dipole resonance occurs due to the opposite flowing currents between the circular patch and ground [25]. The surface current distributions reveal that the proposed unit cell excites both electric and magnetic resonances, with currents flowing in the same and opposite directions, respectively. As a result, the polarization conversion metasurface operates effectively over a wide frequency band.

3.5 Parametric study

To explore the behavior of the proposed metasurface in different frequency bands, we conducted simulations by adjusting the physical dimensions of the meta cell. Figure 3(a) illustrates that the original frequency band of 8-13 GHz can be shifted to a higher frequency range. By decreasing the physical parameters using scaling factors ($sf$) of 0.95, 0.90, and 0.85, we observed frequency band shifts to 8.25-13.85 GHz, 8.75-14.30 GHz, and 9.3-14.8 GHz, respectively, while maintaining a PCR greater than or equal to 90%. Conversely, Fig. 3(b) shows that by increasing the physical parameters in the x-y-plane with $sf$ of 1.1, 1.15, and 1.20, the working frequency band of the PCM can be shifted to a lower frequency range. This resulted in shifted bands of 7.20-12.10 GHz, 6.9-11.60 GHz, and 6.55-10.85 GHz, respectively. Furthermore, Fig. 3(c) demonstrates that by adjusting the radius of the outer split ring (Rp1) in the top metallic patch, the lower-bound frequency of the working band can be shifted. For example, when rp1 is adjusted from 4.68 mm to 4.23 mm, the lower-bound frequency of 7.8 GHz is shifted to 8.25 GHz, while maintaining a PCR greater than or equal to 80 Similarly, Fig. 3(d) shows that by increasing or decreasing the radius (Rp2) of the central circular patch, the upper frequency of the working band can be modified. The upper frequency of 13.2 GHz can be shifted to 12.5 GHz and 14.5 GHz by adjusting Rp2 accordingly.

 

Fig. 3. (a) Variation of the PCR when physical dimensions are scaled by scaling factor $sf$ = 0.85, 0.90, 0.95, 1.0 and (b) $sf$ = 1.0; 1.10; 1.15; 1.20 in xy-plane and when (c) $R_{p1}$ and (d) $R_{p2}$ are varied.

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3.6 Experiment

To experimentally validate the simulated results, a sample of the proposed metasurface was fabricated on a grounded substrate with dimensions of $195 \times 195 mm^2$. The prototype consisted of a $15\times 15$ array of elements etched using standard PCB techniques. Wide-band horn antennas operating in the frequency range of 1.0-18 GHz were employed for irradiating the surface and receiving the reflected waves as depicted in Fig. 4(a). The received signals were measured using a performance network analyzer (PNA-X N5245B). Co-polarized and cross-polarized reflection coefficients were measured by aligning the horn antennas in the same orientation and in orthogonal orientations, respectively. After calibrations were performed for magnitude and phase, the simulated and measured co- and cross-polarization reflection coefficients were obtained when the incident wave was x-polarized as depicted in Fig. 4(b). The simulated and measured PCR exhibited good agreement, as shown in as depicted in Fig. 4(c). Small discrepancies between the simulated and measured results can be attributed to imperfections in the manufacturing process.

 

Fig. 4. (a) Photograph of measuring setup, (b) simulated measured co- and cross polarized reflection coefficients, and (c) PCR.

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In Table 1, a comparison is provided between this work and previous research. This research offers a more comprehensive analysis of the wideband polarization conversion feature compared to previous studies [2,26–34]. Additionally, the proposed polarization conversion metasurface (PCM) exhibits a wider bandwidth than the design in [26]. The proposed structure has a simpler pattern compared to structures in [27,30,33], which rely on multilayer structures, superstrate layers, or complex connections to achieve wide bandwidth properties. The height of the proposed dielectric structure is either equal to or smaller than the structures listed in Table 1, except for [26], where the height is smaller but with a narrower operating bandwidth compared to the design in this work.

Table 1. Performance comparison with previously presented wideband polarization conversion metasurface.^a

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4. Working principles and methods of RCS reduction

The proposed structure achieves reduction in RCS by controlling the reflection phases to create nulls at the boresight and other directions as well. Two methods are employed to design the structure as a low observable platform. For null formation at the boresight, the unit elements are arranged in four regions forming a quad-tile configuration, as depicted in Fig. 5(a), with the corresponding binary coding shown in Fig. 5(b). Additionally, to minimize the impact of incoming energy from other directions, the unit elements are randomly arranged, as illustrated schematically in Fig. 5(c), with the corresponding binary coding displayed in Fig. 5(d).

 

Fig. 5. (a) The rotated cells in four regions and (b) its binary coded configuration, (c) randomly rotated unit cell and (d) its binary coded configuration and simulated results of the unit cell. (e) Phase and (f) magnitude response of cross-polarized reflection coefficient when rotated with its $0^{\circ }$, $90^{\circ }$, $180^{\circ }$ and $270^{\circ }$ respectively, (g) mono-static RCS.

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The proposed structure is divided into four regions, as illustrated in Fig. 5(a) in a schematic view and (b). Each region consists of a $4\times 4$ array of elements. Region I is formed by a $0^{\circ }$ rotation of the metallic split ring, while region II is formed by a $90^{\circ }$ rotation, region III by a $180^{\circ }$ rotation, and region IV by a $270^{\circ }$ rotation. In Fig. 5(b), a "0" represents the metallic split ring rotated by $0^{\circ }$ or $180^{\circ }$, while a "1" represents a rotation of $90^{\circ }$ or $270^{\circ }$. When an incident electromagnetic wave strikes the surface normally, the reflected wave is the superposition of the reflected waves from region I and region II. Let us assume that the reflected electric field from region I is given below

In region II, the metallic split ring is identical to that in region I, but it is rotated by $180^{\circ }$, resulting in a phase difference of $180^{\circ }$ between the waves reflected from region II and region I, as illustrated in Fig. 5(e). However, is same magnitude as region I, as shown in Fig. 5(f). Therefore, the reflected electric field from region II can be expressed as

Similarly, region III has the same phase and magnitude response as region I, as shown in Fig. 5(e) and (f). Therefore, the reflected electric field from region III is the same as that from region I, as given in Eq. (9). Additionally, the reflected electric fields from region IV and region II are also the same. Thus, the total electric field can be expressed as

From Eq. (10), it is evident that due to superposition, the magnitude of the reflected wave becomes too weak to be detected. However, in most frequency points within the working band, the phase difference between the electric fields from region I and region II is not or is close to $180^{\circ }$. Therefore, in such cases, the total electric field can be expressed as

Consider a metal plane with the same dimension as the proposed structure then the total E-field from the metal sheet will be

To keep the reflected power at a low level, for example

After solving Eq. (14), then

Therefore, the phase difference is

From Eq. (16), it is clear that the reflected power can be reduced if the phase difference between region I and II is about $180\pm 30^{\circ }$. The monostatic RCS when unit cells are arranged in four regions, and the RCS of a metal sheet with an equal size to the proposed structure are depicted in Fig. 5(g). For both x- and y-polarized incident waves, the structure of the plane remains the same, resulting in identical RCS values for both polarizations due to the symmetrical configuration. It is observed from Fig. 5(g) that within the working band, energy is mostly suppressed for frequencies between 8 GHz and 13 GHz, and significant RCS reduction is apparent outside the working band.

Additionally, the 2D bistatic RCS is plotted across the incident angle theta in Fig. 6(a) at frequencies of 8.5 GHz and 12 GHz. It can be observed that compared to the response of the metal plane, maximum RCS reduction occurs at an angle of 0 degrees. The bistatic RCS is reduced in other angular regions as well, but it is noted that the proposed structure exhibits an increase in RCS at certain angles compared to the metal plane. This effect is primarily caused by anomalous scattering energy. Furthermore, the 3D scattering patterns under a normal incident wave at resonance frequencies are obtained for a metal plane and the chessboard configuration with four regions, as shown in Fig. 6(c) and (d). In Fig. 6(c), it is evident that the incoming energy is mostly deflected in four different directions due to the arrangement of unit elements in the four regions. The chessboard-like plane with four regions effectively converts the incoming wave into four beams. Conversely, the metasurface with randomly arranged unit cells scatters the incident microwave field’s energy in all directions, as depicted in Fig. 6(d). From Fig. 5(g), it can be concluded that the monostatic RCS reduction achieved using the chessboard configuration with four regions is more significant compared to the random coding method. However, based on the 3D scattering pattern, it can be inferred that the random coding method performs better than the chessboard configuration with four regions.

 

Fig. 6. (a) 2-D scattering patterns, (b) 3-D scattering patterns of metal plane, and (c) chessboard-like plane with four regions and (d) randomly rotated unit cells with normal incidence.

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5. Radiation mode

In this section, we select the same unit cell discussed in the previous section, which functions as a PCM, to demonstrate the concept of a metasurface antenna. The periodic array of these PCM unit cells forms sub-wavelength particles. These particles exhibit minimal response to the magnetic field but display strong resonance in the electrical field. To facilitate the feeding of these radiating patches, we propose a suitable feeding network consisting of T-shaped microstrip feed lines that connect all the patches to a single feed point.

5.1 Design and simulation

To design a high-gain and low RCS metasurface antenna, an additional F4B substrate with a dielectric constant of 2.65 and a thickness of 1 mm is introduced beneath the ground, as depicted in Fig. 7. This additional substrate enables the creation of a feeding network on the backside of the designed PCM structure. The circular patches on the top layer, shown in Fig. 7(a), are connected to the feeding network designed on the backside of the structure. The feeding network consists of T-shaped microstrip feed lines, as illustrated in Fig. 7(b) and (d). All unit cells are connected to a single 50 Ohm feed line. The corporate feeding network comprises a source impedance of $Z_{50} = 50 \Omega$, a characteristic impedance of $Z_{\circ } = 100 \Omega$, and a quarter-wave transformer with an impedance of $Z_{a}(\lambda /4) = 70.71 \Omega$. The characteristic impedance ($Z_{\circ }$) of the microstrip feed patch is designed with respect to the source impedance ($Z_{50}$), given by $Z_{\circ } = n \times Z_{50}$, where $n$ represents the number of twigs emanating from each node. In the proposed structure, $n = 2$ and $Z_{50} = 50 \Omega$, resulting in a characteristic impedance of $Z_{\circ } = 2 \times 50 = 100 \Omega$. Two microstrip lines are matched using a quarter-wave transformer, with an impedance calculated as $Z_{a}(\lambda /4) = \sqrt {Z_{\circ } \times 50} = 70.71 \Omega$. Initially, 16 elements are arranged in a $4\times 4$ matrix, as depicted in the schematic model of Fig. 7(a). Metallic vias with a diameter of 0.7 mm are inserted in the proposed structure. These vias connect the feeding network to the disc-shaped metallic patches on the top layer, which act as radiators when the feeding network is connected to the source signal. In this configuration, the split ring serves as a parasitic radiator. The metasurface radiator was simulated using CST MWS version 2019 with open boundary conditions. The initial $4\times 4$ elements array was extended to an $8\times 8$ elements array, as depicted in Fig. 7(c) and (d). The simulated results for the $4\times 4$ and $8\times 8$ arrays are shown in Fig. 8. In Fig. 8(a), it can be observed that both arrays achieve good impedance matching bandwidth ($S11<-10$ dB) at resonance frequencies of 8.6 GHz and 12.4 GHz. Figure 8(b) and (c) displays the gain and radiation efficiency versus frequency. It can be seen that the gain increases at resonance frequencies for both the $4\times 4$ and $8\times 8$ arrays. At 8.6 GHz, the gain is 8.08 dB for the $4\times 4$ array and 15.1 dB for the $8\times 8$ array, indicating a 7 dB gain increment with the increased number of elements. Similarly, at 12.6 GHz, the gain is 10.7 dB for the $4\times 4$ array and 16.4 dB for the $8\times 8$ array. The radiation efficiency, defined as $\epsilon _r = G/D$, where $G$ is the gain and $D$ is the directivity, is shown in Fig. 8(c). At both resonance frequencies, the radiation efficiency is approximately 95%, as the gain increase at resonance frequencies. The 3D radiation patterns in Fig. 8(d)-(g) indicate that the main lobe is directed towards the normal direction for both the $4\times 4$ and $8\times 8$ arrays. The gain reaches up to 15.1 dB at 8.6 GHz and 16.4 dB at 12.4 GHz, as depicted in Fig. 8(h)-(i). At 8.6 GHz, the 3 dB angular width and side lobe level (SLL) are 17.5 degrees and -8.8 dB, respectively. Similarly, at 12 GHz, the 3 dB angular width and SLL are 16.4 degrees and -4.7 dB, respectively.

 

Fig. 7. The schematic views of (a) $4\times 4$ elements array, (b) the feeding network of (a) and (c) of $8\times 8$ elements array and (d) the feeding network of (c).

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Fig. 8. Comparisonm of $4\times 4$ and $8\times 8$ arrays in (a) reflection coefficient, (b) gain, and (c) efficiency, (d) 3-D radiation of $4\times 4$ elements array at $8.6$ GHz and (e) $12.4$ GHz; (f) 3-D radiation of $8\times 8$ elements array at $8.6$ GHz and (g) $12.4$ GHz and (h) 2-D radiation pattern at $8.6$ GHz (i) $12.4$ GHz.

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5.2 Experimental verification

To experimentally verify the performance of the designed structure as a radiator, a prototype was manufactured on an F4B substrate. The top and bottom layers of the prototype are shown in Fig. 9(a) and (b), respectively. The 3 mm thick and 1 mm thick F4B substrates were pressed together to form a composite substrate using a heat press. The reflection coefficient (S11) was measured using a PNA-X 5245A. The measured and simulated S11 values are shown in Fig. 9(c). It can be observed that good impedance matching bandwidth is achieved at both resonance frequencies. The measured S11 values are slightly better than the simulated ones, which may be attributed to the slight increase in substrate thickness during the hot-pressing process. The radiation efficiency, shown in Fig. 9(c), improves at resonance frequencies and reaches up to 96.50%. The realized peak gain was measured over the matching bandwidth, as depicted in Fig. 9(d). The maximum gain achieved is 15 dB at 8.6 GHz and 16.5 dB at 12.4 GHz.

Table 2 compares important parameters of the designed structure as a radiator in this work with previously published works. The designed structure in this research demonstrates high gain, low RCS, and high efficiency compared to previous works.

 

Fig. 9. A photograph of fabricated prototype (a) top and (b) bottom layer; (c) its simulated and measured reflection coefficient (S11) and efficiency (d) realized gain.

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Table 2. Performance comparison with previously published metasurface antenna

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Table 3 illustrates the novelty of the metamaterial-inspired structure by showcasing multiple functions. Existing multi-functional surfaces were designed to manipulate incident-free space plane waves. However, in this work, two different functions are effectively integrated: polarization conversion of incident free-space waves and EM radiation of the input source signal. To the author’s knowledge, although structures for EM absorption and radiation have been designed and tested in [19,20], these radiating structures contain resistive elements that degrade the performance in terms of gain, directivity, and efficiency.

Table 3. Overall comparison to previously published works as a multifunctional structure

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6. Conclusion

In conclusion, this work introduces a novel integrated structure with combined polarization conversion and EM radiation functions. The structure consists of a split ring and circular patch, enabling the rotation of linearly incident waves to their orthogonal counterparts across a broad frequency range of 8.0-13.5 GHz. The PCR reaches nearly 100% at resonance frequencies of 8.35 GHz, 10.55 GHz, and 12.70 GHz, with multiple resonances confirming its effectiveness. The fabricated prototype is experimentally validated, demonstrating agreement with numerical simulations. The PCM unit is effectively utilized for RCS reduction using two distinct methods i.e., the random coding and quad chessboard configuration methods. The quad chessboard configuration reduces incoming energy at an angle of 0$^\circ$, while the random coding method reduces energy in other direction as well. These techniques contribute to effective RCS reduction. Additionally, the same PCM structure is repurposed for EM radiation applications. An additional feeding network is designed and applied on the backside of the PCM, connecting to all circular patches on the top layer. The resulting radiator operates at resonance frequencies of 8.6 GHz and 12.6 GHz, exhibiting low RCS and high gain characteristics. Experimental verification of the radiator demonstrates good agreement between simulated and measured results.