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Abstract
A modified reflective metasurface, which is constructed by replacing the metal ground of the reflective coding metasurface with a bandpass frequency-selective surface, is proposed. The metasurface has transmission and reduction of radar cross-section characteristics. This allows the metasurface to overcome the drawbacks of conventional realizations, which use lossy materials. The modified metasurface provides high-efficiency transmission in the passband of a frequency-selective surface and broadband reduction of the radar cross section in the rejection band of the frequency-selective surface. Transmission of −0.24 dB was achieved at 4.6 GHz, as well as a −15 dB reduction of radar cross section from 8.5 to 13.5 GHz. This work provides advancements in metasurface applications.
© 2018 Optical Society of America under the terms of the OSA Open Access Publishing Agreement
1. Introduction
The development of new types of material provides novel means to manipulate light [1–4]. It is possible to realize various functionalities, such as filtering [4], negative refraction [5], light trapping [6], and radar cross-section reduction (RCSR) [7–13]. For practical applications, one device having multiple functionalities would be preferable. For example, a device with simultaneous transmission and RCSR characteristics has attracted considerable attention recently. It can be used to design stealthy radomes and systems where mutual interference should be reduced among different subsystems [14–26].
Structures equipped with both high transmission and RCSR characteristics caused by absorbing electromagnetic waves are known as “rasorbers.” Rasorbers can be categorized as two-dimensional (2D) and three-dimensional (3D) rasorbers. The 2D rasorbers are usually sandwich structures of lossy material, lossless substrate, and bandpass frequency-selective surface (FSS). The transmission behavior is realized in the passband of the FSS. In the rejection band, the FSS works as a metal plate, and the entire structure functions as a wideband absorber [27,28], leading to broadband RCSR. The 3D rasorber is based on a 2D periodic array of multimode cavities [24–26]. In this scheme, lossy and lossless resonators are constructed in parallel form, leading to arbitrary absorption bands at two sides of the passband. However, 2D and 3D rasorbers suffer from the drawbacks brought by lossy materials, such as a high loss in transmission [14–17], large density [15–17], soldering of many lumped elements [18–20], and instability in harsh environments [21–23].
In addition to absorption, it is possible to realize broadband RCSR by diffusion. Reflective coding metasurfaces (CMs) [9–13] work in this manner. A reflective CM is the sandwich structure that consists of a CM with n (n = 1, 2, 3 …) bits of different metallic patterns, a dielectric slab, and a metal plate. Different metallic patterns bring different reflective phases, but the gradient of the phase discontinuity is the constant (π/n). For example, binary reflective CMs are composed of two types of basic element with 0 and π reflective phases. When the elements are randomly arranged, destructive interference occurs. The electromagnetic energy shoot on the CMs is spread in various directions, resulting in broadband RCSR. Replacing the metal ground of CMs with bandpass FSSs, it is possible to realize low-loss transmission and wideband RCSR in two distinct frequency bands. For example, Huang et al. proposed such a kind of structure (CM-FSS) and illustrated the qualitative working principles [29]. Because no lossy material is used in CM-FSS, the shortcomings of the rasorbers mentioned above are avoided. Here, the working mechanism and design methods of CM-FSS are complemented with a sandwich structure of flowerlike patterns, dielectric slab, and Jerusalem-cross slot FSS. A balance between the transmission and RCSR functionalities is realized using a particle swarm optimization (PSO) [30] procedure. As a result, the insertion loss at 4.6 GHz is reduced to −0.24 dB, which is smaller than those of rasorbers previously reported [5–17]. In addition, −15 dB RCSR from 8.5 GHz to 13.5 GHz was achieved. Both high-efficiency transmission and broadband RCSR were verified by experiments.
2. Design and discussion
Figure 1 shows a diagram and the working mechanism of the CM-FSS. The metal ground of a reflective CM is replaced with a bandpass FSS. In the diffusion band of the CM-FSS, the bandpass FSS works as a metal plate. The incoming electromagnetic energy is reflected into various directions for the destructive interface among the different elements of the CM. In the passband, both the CM and bandpass FSS show low-loss transmission. The incoming electromagnetic energy passes through the CM-FSS with low loss.
As an example of the above concept, the CM-FSS displayed in Fig. 2 was considered. It consisted of a CM of flower-like metallic patterns, a dielectric slab [εr = 2.6(1 - j0.001)], and a metallic Jerusalem-cross slot FSS. The Jerusalem-cross slot FSS was used because it has a larger rectangle coefficient of transmission compared with other kinds of bandpass FSS [31]. This reduces the frequency spacing between the two functionalities of the CM-FSS. A “superelement” that comprised 2 × 2 basic elements in the metasurface to mimic the periodic boundary condition [12] was adopted. The superelements were randomly distributed to achieve diffusion-like reflection. The details of the basic elements of the CM-FSS are given in Fig. 2. Basic elements are a sandwich structure of a flowerlike pattern, a dielectric slab, and a Jerusalem cross slot.
Fig. 2 Schematic of designed CM-FSS and details of the basic elements [p = 13 mm, t = 4 mm, l = 12 mm, d = 6 mm, g = 0.5 mm, W1 = 0.5 mm, W2 = 0.5 mm, r1 = 1.5 mm, r2 = 2 mm, r3 = 2.5 mm, and r4 = 3.2 mm (Element 1) or 4 mm (Element 2)].
2.1 Broadband RCSR
Before investigating the CM-FSS, the electromagnetic response of the reflective CM based on flower-like patterns was studied. A reflective CM whose parameters are same as those in Fig. 2 was considered. All the parameters were fixed except the arm length of the flower pattern, r4. By adjusting r4, the resonant frequencies of the basic elements of the CM can be tuned, and this will result in different reflection phases.
The reflection coefficients of basic elements were obtained using the commercial software CST Microwave Studio. Figures. 3(a) and 3(c) plot the reflection coefficients of a periodic array of two basic elements. The reflection amplitudes keeps near unity when r4 is changed. The reflection phases are roughly parallel to each other from 7 to 14 GHz; their difference is almost a constant. For clarity, only two reflection phases were plotted. Actually, there are many pairs of reflection coefficients that can be realized using different sizes. To construct a reflective CM, n (n = 1, 2, 3 …) bits of elements can be selected. The RCSR is given by
where pi/∑pi = 1/N is the filling percentage of each element, and ai and φi are the amplitude and phase of the reflection coefficient, respectively. For simplicity, only two elements, whose reflection coefficients are plotted in Figs. 3(a) and 3(c), were used. The filling percentages were 0.5 in this case. In practical applications, reflective CMs are required to have certain RCSR. For example, a −15 dB RCSR requires that the phase difference between different elements appears in the range of 180° ± 20.5°—the gray region in Fig. 3(c). Therefore, the calculated −15 dB RCSR of the reflective CM is 7.2–14 GHz.Fig. 3 Reflection (a) amplitudes and (c) phases of basic elements of reflective CM and reflection (b) amplitudes and (d) phases of basic elements of CM-FSS with different r4; calculated RCSR of (e) reflective CM and (f) CM-FSS.
When the metal plate of the reflective CM is replaced by a metallic Jerusalem cross slot FSS, it is necessary to examine whether the FSS supports the functionality of broadband RCS reduction. The reflection coefficients of a 4-mm-thick dielectric slab [εr = 2.6(1 − j0.001)] backed by a metal plate and a Jerusalem-cross slot FSS were compared. The geometric parameters of FSS are given in the caption of Fig. 2. As shown in Fig. 4, the reflection amplitude of a 4-mm-thick dielectric slab backed by a Jerusalem cross slot FSS is greater than 0.96 from 8.5 to 13.5 GHz. This means most of the incident electromagnetic energy is reflected back. Meanwhile, the reflection phase of the dielectric slab backed by FSS decreases linearly with frequency, which is similar to the case of a metal plate. Furthermore, the phase difference between the two cases is small. Therefore, the functionality of RCSR is still accessible when the metal plate is replaced by a Jerusalem cross slot FSS. This can be verified by Figs. 3(b) and 3(d).
Fig. 4 Reflection coefficients of a 4-mm-thick dielectric slab [εr = 2.6(1 – j0.001)] backed by a metal plate or Jerusalem cross slot FSS.
Because the reflection amplitudes are no longer unities, the CM-FSS structure may change the frequency range for RCSR. According to Eq. (1), the filling percentages of the basic elements are no longer the same. Moreover, the frequency range with a large phase difference shrinks significantly. As a result, the bandwidth of −15 dB RCSR shrinks. How to select the basic elements and the filling percentages of the basic elements is discussed in Section 2.3. This is because the composition of the CM-FSS also influences the transmission performance.
2.2 High-efficiency transmission
To achieve high-efficiency transmission, the flower-like patterns should be transparent at the passband of the Jerusalem cross slot FSS. The flower-like patterns are a type of bandstop structure. It is possible to tune the passband by adjusting the arm length of the flower pattern, r4. Figure 5(a) plots the transmission amplitudes of flowerlike patterns backed by a 4-mm-thick dielectric slab [εr = 2.6(1 − j0.001)]. The transmission of the Jerusalem-cross slot FSS with dielectric slab is also given for comparison. The flower-like patterns show high-efficiency transmission at the passband of the FSS. As a result, the CM-FSS shows high-efficiency transmission in the passband of the FSS.
Fig. 5 (a) Transmission amplitudes of flowerlike patterns and the FSS; transmission (b) amplitudes and (c) phases of basic elements of CM-FSS with different r4.
To evaluate the transmission performance of the CM-FSS, the transmission coefficients of basic elements should be derived. Considering an array of a single basic element, the transmission coefficient can be evaluated as follows [12].
where Z0 is the wave impedance of air, and terms A, B, C, and D are the elements of the transmission line matrix of basic elements, which is evaluated as the product of the three cascaded matriceswhere βd and Zd are the propagation constant and wave impedance of the dielectric slab, respectively. If the geometric parameters of the flowerlike pattern are changed, Zf will also change, leading to the variety of transmission coefficients.Figures 5(b) and 5(c) plot the transmission coefficients of basic elements with different r4. The differences between the transmission coefficients of basic elements in the passband of the FSS cannot be ignored. As a result, part of the transmitted energy is diffused to other directions. The more the types of element used, the stronger the diffusion. As a result, fewer elements should be used to ensure high-efficiency transmission. However, a better broadband RCSR is obtained using flowerlike patterns with different dimensions. Therefore, a tradeoff must be made between the two functionalities: the high-efficiency transmission and the broadband RCSR.
2.3 Optimization and discussion
The passband and diffusion band of CM-FSS can be tuned by varying the geometric sizes of the FSS and the permittivity of the dielectric slab. As an example, a 4-mm-thick polytetrafluoroethylene (PTEE) board [F4BM, εr = 2.6(1 − j0.001)] was used as the spacer between flowerlike patterns and the bandpass FSS. The passband behavior was set at approximately 4.5 GHz and the diffusion band in X band. The CM was set to be composed of two types of element to guarantee high-efficiency transmission. In this case, Eq. (1) can be simplified as
where p1 and (1 − p1) are the filling percentages of the two types of element, and ari and φri are the reflection amplitude and phase of each element, respectively. Similarly, the transmission reduction is expressed aswhere ati and φti are the transmission amplitude and phase of each element, respectively.The key point in designing the CM-FSS is to achieve a balance between the two functionalities according to actual needs. Suitable geometric dimensions of basic elements and filling percentages were selected to construct a CM-FSS. The reflection and transmission coefficients of the basic elements were calculated using numerical methods, and a database of the coefficients was made by varying the geometric sizes of the flowerlike patterns. Next, the PSO method was used to optimize the composition of the CM-FSS. In the optimization, the minimum insertion loss was guaranteed to be smaller than 0.5 dB. The performance of the CM-FSS was evaluated by a fitness function, which requires that the relative −15 dB bandwidth of RCSR goes to the maximum
where frequp and freqdown are the highest and lowest frequencies that satisfy the condition.The optimized parameters of the CM-FSS are given in the caption of Fig. 3. Their filling percentages are 0.51 and 0.49, respectively. As stated above, the frequency range with a large phase difference among the reflection phases shrinks. As shown in Fig. 6, the calculated −15 dB bandwidth of RCSR is 8.5−13.5 GHz, which is smaller than that of the reflective CM (7.2−14 GHz). The transmission coefficients of the two elements are plotted in Fig. 5. The differences of transmission amplitudes and phases are very small in the passband. High-efficiency transmission was realized with the optimized CM-FSS. The optimized bandwidth of −1 dB transmission was 3.9−4.8 GHz. Moreover, the insertion loss of transmission was minimized to −0.5 dB at 4.6 GHz.
To have a better understanding of the mechanism of the design, the energy distribution when the CM-FSS is illuminated by a plane wave was plotted. The diffusion consisted of two parts: the energy diffused backward (∑piari2 − RCSR) and the energy diffused forward (∑piati2 − TR). The absorption was the remaining part of the energy (1 − ∑piari2 − ∑piati2). As shown in Fig. 6, the proportion of absorption is very small, resulting in high-efficiency transmission at the passband of the FSS. In addition, only a small fraction of the energy goes through the CM-FSS in the diffusion band. Most of the energy is diffused in various directions. Furthermore, the electric field distribution of Jerusalem-cross slot FSS and flower-like patterns are plotted. At the transmission band [Figs. 7(a) and (b)], the electric filed in the slots of FSS and flower-like patterns are quite strong. The electromagnetic energy goes through the CM-FSS. At the diffusion band [Figs. 7(c)–6(f)], the electric filed in the slots of FSS become much weaker while the electric filed on the flower-like patterns become much stronger. This just verify that the energy is reflected backward.
Fig. 7 Simulated electric filed distribution of Jerusalem-cross slot FSS [(a), (c) and (e)] and flower-like patterns [(b), (d) and (f)] at 4.6 GHz [(a) and (d)], 8.7 GHz [(b) and (c)] and 13.3 GHz [(e) and (f)].
3. Measurements
As a validation, a prototype (300 mm × 300 mm) comprising 20 × 20 super-elements was fabricated, and it is shown in Figs. 8(a) and 8(b). The flower-like metallic patterns and FSS were printed on two sides of a 4-mm-thick PTEE board [F4BM, εr = 2.6(1 −j0.001)]. The performance of RCSR was tested in a microwave anechoic chamber. As shown in Fig. 8(c), the measured −15 dB RCSR covers 8.5−13.5 GHz. The results agree well with the calculated one. Besides, we measured the angular dependence of the RCSR, which is important for practical applications. The results are shown in Figs. 8(e) and 8(f). For TE polarization, the RCSR is with small change in the −10 dB bandwidth when the incident angle is smaller than 20°. For TM polarization, the −10 dB bandwidth does not change much up to the incident angle θ = 30°.The transmission performance of the prototype was tested with a free space method. The measured insertion loss at 4.6 GHz was only −0.24 dB, and the −1 dB transmission covers 4.1−5.0 GHz, which agrees well with the results calculated by Eq. (5). The insertion loss was smaller than for a structure proposed previously [1–11]. The results show that combining a CM and FSS can bring more efficient transmission.
Fig. 8 (a) Top and (b) bottom view of the fabricated sample; calculated and measured (c) transmission and (d) RCSR of proposed CM-FSS; RCSR under oblique incidence for (e) TE polarization and (e) TM polarization.
In the work of Huang et al., the two bits of metasurface are arranged in a way of chess board. Most of the energy wave is reflected along the four main directions. In our design, the bits are arranged randomly. The energy is reflected along various directions. As a result, our structure is more difficult to detect. Furthermore, we realize a balance between the transmission and RCSR functionalities using a PSO procedure. The insertion loss at 4.6 GHz is reduced to −0.24 dB, which is smaller than that of Huang’s structure. However, more work needs to be done to improve the angle stability of functionality of RCSR.
4. Conclusions
A CM and bandpass frequency selective surface were combined to achieve high-efficiency transmission and broadband RCSR simultaneously. Because no lossy material is used in in the design, the proposed structure avoids the shortcomings of rasorbers. With the help of PSO, a balanced performance of the two functionalities was realized. The insertion loss was minimized to 0.24 dB, and the −15 dB RCS reduction covered 8.5−13.5 GHz. The performance of the structure was validated by experiments. With the advantages of easy preparation and lower insertion loss, the structure is more suitable for practical applications than others that have been proposed. This study makes advancements in the application of metasurfaces.
Funding
National Natural Science Foundation of China (NSFC Grant Nos. 61771237; 61671232).
Acknowledgments
This work is partially supported by the Project funded by the Priority Academic Program Development of Jiangsu Higher Education Institutions, and Jiangsu Provincial Key Laboratory of Advanced Manipulating Technique of Electromagnetic Waves.
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