Abstract: We report initially the design, fabrication and measurement of using waveguided electric metamaterials (MTM) in the design of closely-spaced microtrip antenna arrays with mutual coupling reduction. The complementary spiral ring resonators (CSRs) which exhibit single negative resonant permittivity around 3.5GHz are used as the basic electric MTM element. For verification, two CSRs with two and three concentric rings are considered, respectively. By properly arranging these well engineered waveguided MTMs between two H-plane coupled patch antennas, both numerical and measured results indicate that more than 8.4dB mutual coupling reduction is obtained. The mechanism has been studied from a physical insight. The electric MTM element is electrically small, enabling the resultant antenna array to exhibit a small separation (λo/8 at the operating wavelength) and thus a high directivity. The proposed strategy opens an avenue to new types of antenna with super performances and can be generalized for other electric resonators.
©2012 Optical Society of America
In recent years, electromagnetic (EM) metamaterials (MTMs) have spurred a renewed interest in physics and engineering communities due to their abundant abnormal characteristics that are not occurring in nature. The initial impetus for the MTM research was driven by the left handed (LH) MTMs with negative refractive index . Since the MTMs are proposed, theoretical characterized and experimentally realized [2–4], most of interest and explosive works have been directed toward pushing these specific behaviors into engineering applications such as super lens [5, 6], cloaking [7, 8], absorbers [9, 10], antennas [11–13], and other related devices [14, 15]. Despite these fruitful developments, there was still insufficient progress toward practical applications .
Nowadays, reducing antenna size and the resultant mutual coupling is a challenging, common and contradictive issue in communication industry. It has been well established that antenna elements should be spaced at least by λ/2 in order to obtain a decent isolation between the antenna elements, and thus in turn good antenna performances. Otherwise, undesirable degradation on antenna directivity would be encountered due to the near-field effects. However, this requirement certainly conflicts with current increasing demand on antenna miniaturization. A survey of literature indicates that several research efforts have been devoted to mitigating the mutual correlation between coupled antennas, e.g., by using single-negative (SNG) magnetic metamaterials [17, 18], mushroom or uniplanar planar EM band-gap structures (EBG) [19, 20], and by introducing the resonant defects or slits in the ground [21, 22]. Although a fairly good isolation between antenna elements has been achieved, most of the insulating structures are still large, resulting in a big element separation. Moreover, all the utilized MTM structures are confined to the magnetic one, making an alternative and improved schematic a pressing task.
In this paper, we propose a novel strategy to reduce both the mutual coupling and element separation enlightened by the concept of a waveguided MTM (WG-MTM) . Distinguishing from that a lot, this implementation utilizes complementary spiral resonators (CSRs)  which is a fundamental building block of SNG electric MTMs. Moreover, the electric MTM particle is electrically smaller, advancing a step further in homogenized concept. As a consequence, when the electric WG-MTMs with the insulation region coving the operation frequency of antenna elements are incorporated, a comparable isolation between the resultant arrays is expected even in a more closely distributed separation relative to previously reported implementations. Therefore, the proposed strategy is capable of combating the not well-addressed issue and thus can be popularized in real practical applications.
2. Design and characterization of a SNG electric WG-MTMs
WG-MTM is a special kind of artificial materials residing in the planar waveguide environment  which commonly consists of two parallel metallic plates: the top complete metal and the bottom defected one. Enlightened by this concept, Fig. 1 depicts the schematic of the proposed electric WG-MTM element as well as the simulation setup by incorporating square CSRs. To not lose generality, the CSRs geometry can be arbitrary, e.g., circle, hexagon, and octagon and even fractals and are selected as square in this paper for design convenience. Note that it is preferable to select fractal geometry for a smaller unit cell, however, the design procedure would be complicated. To ease the design, the periodicity p is also chosen as square. Similarly, the resultant element is composed of three layers: the top metallic layer, the bottom defected metallic layer on which CSRs are etched, and the middle substrate layer which supports the two metallic plates. This configuration renders an easy fabrication compatible with available printed circuit board (PCB) technique. For a comprehensive study, two cases are considered. The former is the CSRs with two interconnected complementary rings, whereas the latter is that with three interconnected rings. They are denoted as CSR1 and CSR2, respectively for convenience. Note that the CSRs are selected instead of complementary split ring resonator (CSRRs) from the point of view of an electrically smaller particle. This is because that the inductances (L) of CSR1 and CSR2 with low symmetry are four and eight times that of CSRRs by applying the duality principle to the resonant magnetic SRR structures .
For characterization and design, the commercial full-wave finite element method (FEM) EM field simulator Ansoft HFSS is employed and the entire circuits including the upcoming antennas are built on a commonly utilized F4B substrate with a dielectric constant εr = 2.65, thickness h = 1.5mm and loss tangent tanδ = 0.001. In the simulation setup shown in Fig. 1(b), the waveguide is illuminated by a dominant transverse electric and magnetic (TEM) plane wave with wave vector along z-direction and electric field (E-field) polarized along x-direction. The CSRs will response to the axial time-varying E-field components and in turn render an electric response. Two wave ports are distributed at the front and back side far away from the CSRs structure to avoid undesirable high-order modes and near-field cross coupling between the ports and the CSRs. The two walls along y-axes are assigned as perfect magnetic conductors (PMC) to mimic an infinite array. Due to the radiation of the CSRs slot, a sufficient larger air box (with height of 4mm) whose bottom surface is assigned as radiation boundary is continuing the substrate layer to prevent reflections from the computational domain. In this circumstance, the unit cell together with its metallic plates, and the volume between them constitutive the effective medium whose effective permittivity and permeability can be extracted by using standard retrieval procedure . In this process, the phase reference planes are chosen to be just on both sides of the periodicity.
Figure 2 plots the simulated scattering parameters and the retrieved constitutive EM parameters against frequency. Note that these correlative results are provided under several cases such as different main geometrical parameters, various gap orientations and number of unit cells N in the longitudinal direction, aiming at providing a systematic characterization and useful guidelines in antenna design. Referring to Fig. 2(a), very obvious band-gaps identified from the transmission dips are ambiguously observed in all cases. The dip in the transmission coefficients is attributable to the electric resonance of the developed particle when impinges to the axial E-field of the incoming wave. As a result, a single negative permittivity is expected in Fig. 2(d). A further inspection indicates that the frequency (f0) of the resonant dip can be modulated by the geometrical dimensions, e.g., f0 increases when a shrinks and d2 increases. The enhancement of d2 equals to the reduction of the inner ring and in turn the shortened current path in the ground. Remark that the residual parameters such as d1 and g1 (the influence are not given for brevity) play a negligible role in affecting f0. Moreover, the fractional bandwidth (FBW) identified by the isolation better than 10dB is drastically increased from 90 to 240MHz when N increases from one to two. This can be further validated from the CSR2 case shown in Fig. 2(c), where the FBW also drastically enhances from 50 to 320MHz when N extends from one to three. The resultant wider isolation bandwidth is induced by the coupling effect generated between adjacent cells which enhances when more cells are cascaded. In both cases, the band-gap property is insensitive to the gap orientation except for a litter better suppression when the gap is rotated by 90°. Following Fig. 2(b), it can be learned that the periodicity has a negligible effect on the insulating response provided that the dimensions of CSR are fixed.
3. Applications to microstrip antenna array
Mutual coupling is a long-hold but a very essential issue in the antenna discipline. It occurs in terms of both surface waves and space (radiated) waves which deteriorate the antenna performances in several ways such as the grating lobes generation, scan blindness, input impedance mismatch, beam shape degradation and high side-lobe level.
Based on the aforementioned results and guidelines, the exhibited band-gap characteristics of the SNG electric MTMs in the waveguide environment can be directly employed to decouple two closely-placed microstrip patch antennas. To illustrate this capability, two microstrip antenna arrays using the proposed CSR1- and CSR2-loaded electric WG-MTMs are designed at 3.5GHz (Wimax), respectively. The parameters of the CSRs structures are carefully designed in a way that the engineered band gap is able to accommodate the working frequency of the antenna. Figure 3 depicts the schematics of these antenna arrays. As is shown, the proposed antenna arrays are composed of two microstrip patches sandwiched with a bulk of the periodically arranged WG-MTMs. The top metallic plate of the waveguide is aligned with the patch layer, whereas the metallic plate with patterned CSR1 and CSR2 in the bottom metallization of WG-MTM functions as the ground plane of the antenna array. Hence, the loading process of WG-MTM is completely compatible with the fabrication process of the microstrip antennas.
In the former case, a bulk MTM extending to 7 × 2 of the CSR1-loaded WG-MTM element is embedded between two microstrip patches, whereas in the residual case a 9 × 3 array of CSR2-loaded WG-MTM element is inserted between microstrip patches with identical dimensions. Since the operating frequency and signal inhibition level are insensitive to the periodicity once the CSRs layout is fixed, see Fig. 2(b), the edge-to-edge distance ls between these antenna arrays is mainly dependent on the decoupling efficiency and partially on the dimensions of the WG-MTM unit cell. Taking the advantage of compact nature and high decoupling efficiency of the CSRs, ls can be engineered only a small portion of the wavelength at 3.5GHz (λo/8.08 and λo/7.5, respectively). In both cases, the whole footprint occupies a PCB area of 111.4 × 55.4mm2. For characterization, the S-parameters and the antenna radiation patterns are computed using HFSS. Moreover, the conventional patch antenna without SNG electric WG-MTM is also characterized and eventually fabricated and measured for comparisons.
4. Results and discussions
For experimental use, the designed two MTM antennas and the conventional reference antenna are fabricated and measured. To make fair comparisons, the reference antenna occupies the same area of the patch and ground plane, and is built on the same substrate with the proposed MTM antennas. Figure 4 depicts the photograph of the fabricated prototypes. For verification, the S-parameters are characterized like two-port devices through a N5230C vector network analyzer (see the measurement setup in Fig. 5 ), whereas the radiation patterns are through a far-field measurement system in an anechoic chamber by exciting one port of the antennas and loading the other with a 50Ω broadband load.
Figure 5 compares simulated and measured S-parameters against frequency for the proposed WG-MTM antennas and reference antenna. Very consistent results are observed in all cases except for a slight frequency shift upwards (with maximum 0.1GHz) in the measurement condition, indicating the effectiveness of the design. These deviations are mainly attributable to the unstable dielectric constant of the substrate and the infinite CSRs arrays (two and three rows in practice) along y-axis in the simulation, and partially to the tolerances that are inherent in the fabrication process. Nevertheless, the discrepancies are within an applicable tolerable range and have nothing to do with the illustration of the coupling reduction. As is appreciated, the coupling for the H-plane coupled patches has been significantly diminished from the coupling coefficients S21, e.g., the peak S21 is on the order of -11.66dB in the case of conventional antenna, whereas it is drastically reduced to -20.02 and -22.86dB for the CSR1, and CSR2-loaded MTM antenna, respectively. Therefore, a mutual coupling reduction of 8.36 and 11.2dB compared to the reference antenna has been realized for the proposed two MTM antennas. These levels of coupling suppression, which are one of the best results to the authors’ best knowledge, are very desirable and comparable for antennas placed in such a close proximity and thus are of particular interest in practical applications.
To examine the band-gap behavior on the antenna radiation performances, Fig. 6 shows the far-field radiation patterns. Notice that the antenna performances are evaluated under the condition of one-port excitation (element radiation behavior not array behavior). Following the figure, it is observed that all three samples show similar radiation characteristics, including the forward gain, antenna’s efficiency and cross-polarization level in the two principle radiation planes (E-plane and H-plane). Moreover, the CSR arrays help in achieving an improved front-to-back ratio of the patterns, namely reducing the back radiation. Despite this, measurement results do not show any additional significant differences and notable changes at the operating frequency by loading the WG-MTMs. Therefore, it is expected that the antenna array performances would be significantly improved when each antenna element is fed due to the suppressed element mutual coupling and the maintained element radiation performances.
To provide a physical insight into the working mechanism of the isolation, Fig. 7 illustrates the numerically calculated surface current distributions on the top patch in three cases. The currents are obtained from HFSS by exciting one antenna while terminating the other with a standard 50Ω impedance matching load. Following the Fig., one can clearly find that a high concentration of the surface currents is observed in the three excited antennas. Without the CSRs, strong currents are coupled from the excited antenna to the adjacent one. In contrast, the electric power is mostly trapped and absorbed around the interface between the WG-MTM and excited patch and then is progressively attenuated within the WG-MTM array, enabling most blocked radiated energy and mitigated transmitted energy in adjacent patch. The almost consumed power has unambiguously demonstrated the effectiveness of the WG-MTM as an antenna decoupler. This is especially true for the MTM antenna using three rows of CSR2 which is more efficient in maintaining the low correlations between antenna elements.
We have designed, fabricated and characterized a type of antennna arrays by using CSRs-loaded SNG elctric WG-MTMs. The resultant antennas advance in many aspects such as more than 8.4dB reduction in mutual coupling, 0.37λo reduction in size and easy fabrications without any deterioration in the far-field properties compared to the initial array structure. Unlike EBG and previous SNG magnetic MTM-based coupling reduction strategies, the utilized insulating structures are very compact which is of a reference value in practical applications, and no additional metallic structures are required to embed inside the substrate. Most importantly, the proposed method can be popularized in a set of other electric resonators, opening a way in new types of antennas with super performances.
This work is supported by the National Natural Science Foundation of China under Grant No. 60971118 and the 973 Project of Science and Technology Ministry under Grant Nos.2009CB613306, and is also supported by the Innovation Foundation for Postgraduate’s Dissertation of Air Force Engineering University under Grant DY12101.
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