Quadrature phase-shift keying (QPSK) is usually generated using an in-phase/quadrature (IQ) modulator in a balanced driving-condition, showing a square-shape constellation in complex plane. This conventional QPSK is referred to as square QPSK (S-QPSK) in this paper. On the other hand, when an IQ modulator is driven in an un-balanced manner with different amplitudes in in-phase (I) and quadrature (Q) branches, a rectangular QPSK (R-QPSK) could be synthesized. The concept of R-QPSK is proposed for the first time and applied to optical eight-ary phase-shift keying (8PSK) transmitter. By cascading an S-QPSK and an R-QPSK, an optical 8PSK could be synthesized. The transmitter configuration is based on two cascaded IQ modulators, which also could be used to generate other advanced multi-level formats like quadrature amplitude modulation (QAM) when different driving and bias conditions are applied. Therefore, the proposed transmitter structure has potential to be deployed as a versatile transmitter for synthesis of several different multi-level modulation formats for the future dynamic optical networks. A 30-Gb/s optical 8PSK is experimentally demonstrated using the proposed solution.
©2011 Optical Society of America
Advanced optical multi-level modulations formats, such as quadrature phase-shift keying (QPSK), eight-ary phase-shift keying (8PSK), and quadrature amplitude modulation (QAM), become promising to meet the increasing demand on spectral efficiency and high throughout in optical communication systems. Among existing multi-level formats, 8PSK becomes attractive for its high tolerance against fiber nonlinearities, moderate implementation complexity, and flexibility for de-modulation in either coherent approaches [1–6] or direct detection [7-8]. Several schemes have been reported to generate optical 8PSK, including: i) a cascade of a partially-driven (π/4-shift or π/2-shift) phase modulator (PM) with in-phase/quadrature (IQ) modulator, or Mach-Zehnder modulator (MZM) [1–3,7,8]; ii) high-order integrated quad-parallel MZM ; iii) two cascaded QPSK modulators with an interferometer in between , iv) single IQ modulator with multilevel electrical signals [6,7]. However, these techniques usually suffer from one of the following drawbacks: i) the deployment of PM may introduce extra phase chirp, degrading the tolerance against dispersion and the modulation bandwidth limitations, ii) implementation complexity and cost are relatively high for the highly-integrated modulator; iii) extra insertion loss is introduced due to the use of an interferometer in the transmitter side; iv) the performance requirement for both the modulator and electrical drivers is high when handling multi-level electrical signals.
Recently, we have proposed and successfully demonstrated an optical 16-ary QAM transmitter using two tandem IQ modulators . In this paper, we demonstrate that an optical 8PSK could also be synthesized using the same optical frontend. In other words, the transmitter could be configured to generate different optical multi-level formats. Therefore, it offers the flexibility in the modulation formats for adapting to the dynamics in future optical networks. Moreover, compared with the 8PSK transmitter schemes using PM, the generated optical 8PSK using the proposed scheme has superior tolerance against bandwidth limitation, and less phase chirp, thus offering higher dispersion tolerance.
2. Operation principle
Usually QPSK is generated using an IQ modulator in a balanced driving-condition, resulting in the square-shaped constellation in complex plane (referred to as S-QPSK in this paper). When an IQ modulator is driven in an unbalanced manner, a rectangular constellation could be obtained (referred to as R-QPSK in this paper). By cascading S-QPSK and R-QPSK, an optical 8PSK could be synthesized from two tandem IQ modulators under different driving conditions. The operation principle of the proposed R-QPSK and 8PSK transmitter is illustrated in Fig. 1 . Two pairs of independent data streams, (D1, D2) and (D3, D4), are applied to two cascaded IQ modulators (IQ-1 and IQ-2) to generate R-QPSK and S-QPSK, respectively. In IQ-1, unbalanced driving electrical signals are applied to in-phase (I) and quadrature (Q) branches to generate R-QPSK. One of the sub MZMs in IQ-1, e.g. MZM-I, is under-driven by a peak-to-peak voltage of 0.8Vπ and biased at the null point, whereas the other one, e.g. MZM-Q, is fully-driven by 2Vπ. Both of these two embedded sub MZMs are biased at the null point of the transmission curve. This unbalanced configuration would result in the unequal amplitudes between I and Q branches with an amplitude ratio a:b of ~0.4, thus generating a R-QPSK constellation in complex plane. The amplitude ratio between I and Q branches could be finely tuned by properly adjusting the driving voltage in the sub-MZM of I branch. The relative phase angle between the four phase symbols is 45 or 135 degree. The other IQ modulator (IQ-2) is configured as a standard QPSK transmitter, generating four phase states in a symmetric square shape, i.e. S-QPSK. In the S-QPSK constellation, the same amplitudes are obtained for the generated symbols in I and Q branches of IQ-2, i.e. amplitude ratio c:d = 1. Four phase rotations, i.e. 45°, 225°, 135° and 315°, are introduced for the incoming R-QPSK generated in IQ-1. Four symbols in different shapes are used to denote the phase states generated in IQ-2. Because of the symmetric features of both R-QPSK and S-QPSK, two sets of R-QPSK symbols overlap each other (45° and 225°, 135° and 315°) in the resultant phase pattern after the two tandem IQ modulators. Therefore, the S-QPSK actually provides a binary 90° phase rotation, although four phase states are generated in IQ-2. Assuming gray coding is applied, the 90° phase rotation is logically determined by the result of XOR operation between the two driving sequences of the S-QPSK, i.e. D3⊗D4. Thus, the symbol definition in the obtained optical 8PSK is logically determined by binary driving steams through (D1, D2, D3⊗D4). The symbol definitions of the generated R-QPSK, S-QPSK and optical 8PSK are illustrated in Fig. 1 as well. The required coding is relatively simple, compared with those deployed in other reported schemes like [4, 8]. Besides, since only conventional IQ modulators are deployed in the 8PSK transmitter, compared with the scheme based on high-order integrated modulator , it is relatively easier to offer stable performance against temperature fluctuation by simply deploying commercially-available bias controllers.
Figure 2(a) shows a complex-envelope plot of 8PSK obtained from the proposed transmitter, which consists of two cascaded IQ modulators (known as 2-IQ scheme here). For comparison, Fig. 2(b) plots another 8PSK constellation generated using three PMs in series (referred to as 3-PM) . The same parameters such as the bandwidth of deployed electrical drivers and optical modulators were applied for these two schemes. To assess the constellation and phase transition, the optical field is plotted directly after the modulator without introducing any noise. In the ideal case where rectangular impulse electronics are applied, these two transmitters would produce the same output with no difference in the phase transition. When low-pass Gaussian filtering with a cut-off frequency of 0.6 times the symbol rate is applied at the driving electronics, different transition patterns among symbols are obtained for these two transmitter schemes. It hence results in different constellations, which in turn gives different system performances such as bit-error rate (BER) and tolerance against dispersion or electrical bandwidth. Note that, the inter-symbol interference (ISI) mentioned in this paper is merely caused by the finite frequency bandwidth in the transmitter side. Obviously, for the scheme 3-PM, the symbol clouds are more spread and dislocated than the scheme 2-IQ due to the narrowed electronic bandwidth. Therefore, it is clear that the scheme 3-PM is more sensitive to the bandwidth limitation occurred in both electrical drivers and optical modulators. Besides, strongly-curved phase transitions are observed in the scheme deploying PM (3-PM), indicating that less phase chirp is obtained in the 2-IQ scheme. It then results in superior dispersion tolerance. More comprehensive performance comparison among these reported 8PSK transmitters are now under investigation.
3. Experiment and results
An experiment at 10 Gbaud was carried out to verify the proposed scheme. Figure 3 illustrates the schematic diagram of the experimental setup. Light from an external cavity laser (ECL) with around 100-kHz line-width was fed into the proposed 8PSK transmitter, which consists of two LiNbO3 IQ modulators (IQ-1 and IQ-2) in series. The deployed IQ modulator has a 3-dB electro-optic modulation bandwidth of around 16 GHz, and a half-wave voltage (Vπ) of around 5.5 V. Four binary electrical signals (D1~D4) with pseudorandom binary sequence (PRBS) 215-1 were applied to drive these two IQ modulators. To obtain an R-QPSK, the two embedded sub-MZM in the modulator IQ-1 were driven in an unbalanced manner by two electrical signals (D1, D2) with different peak-to-peak voltages of around 4 V and 11 V, respectively. The following IQ modulator, IQ-2, was fully-driven by the other two streams (D3, D4) to generate an S-QPSK phase pattern. An erbium-doped fiber amplifier (EDFA) and a tunable optical delay line were inserted between the two IQ modulators for maintaining a high OSNR and time-domain synchronization. To avoid device damage or performance degradation in the transmitter, the output power of EDFA was set at less than 8 dBm. In order to maintain the correct timing between the cascaded IQ modulators over time, we can simply deploy a feedback control system, which is similar to the technology typically used in two-stage return-to-zero transmitters for synchronizing pulse-carver and data modulator . At the receiver side, the generated 8PSK was demodulated by using a coherent phase-diversity digital receiver, which includes a local oscillator (LO), an optical 90° hybrid, two pairs of balanced-detectors, high-speed analog-to-digital converters (ADCs) and offline digital signal processing (DSP) unit. A polarization- and phase-diversity coherent receiver could be deployed, to fully solve the polarization-dependent issue in the receiver. The LO also has a narrow line-width of around 100 kHz with a <500 MHz frequency offset from input 8PSK signal. The power of LO and input signal was set at around 9 dBm and −3 dBm, respectively. After optical-to-electrical (OE) conversion, the signal was then sampled for A/D conversion using a real-time oscilloscope (sampling rate: 50 Gsamples/s; analog bandwidth: 12.5 GHz; resolution: 8 bits). The captured data was then processed offline by DSP. It performs several functions, such as clock recovery, oversampling, retiming, the carrier-phase estimation, FIR filtering and so on. I and Q components were finally recovered for constellation reconstruction and BER estimation.
The recovered R-QPSK and S-QPSK constellations are shown in Fig. 4(a) and (b) , respectively. As shown in Fig. 4(c), an optical 8PSK constellation was synthesized when both of the IQ modulators were activated. Instead of coherent detection, it is also possible to directly detect 8PSK after delay-interferometer. The captured eye diagram after direct-detection is depicted in Fig. 5 (a) . The clear eye-opening was observed after direct detection. The eye diagram of 8PSK just after the transmitter is also presented in Fig. 5(b), showing a constant envelope.
By adjusting the optical signal-to-noise ratio (OSNR, 0.1 nm) of input signal at the coherent receiver, the BER of 30-Gb/s optical 8PSK was evaluated using offline processing. The theoretical and measured BERs as function of receiver OSNR are depicted in Fig. 6 . Around 40 000 symbols were used for BER counting, corresponding to the evaluable minimum BER of around 8x10−6. Compared with the theoretical value, around 7-dB power penalty was found at BER of 10−3. The reasons for the power penalty include: i) The equalization and decision technology used in the coherent receiver was not optimized for 8PSK; ii) Since one of the IQ modulators was not fully driven, the system performance was much easier to be affected by the impairments in driving electronics. The BER performance could be further improved by optimizing the equalizer and decision boundaries  for 8PSK in the deployed coherent receiver, and deploying specially-designed unbalanced IQ modulator for generating R-QPSK. Around 20-GHz-wide main lobe is observed in the measured optical spectrum (Fig. 7 ). It confirms that the optical bandwidth of 8PSK is similar to that of binary or quadrature phase-shift keying at the same symbol rate.
In this paper, we have introduced the R-QPSK for generating optical 8PSK, which has not been proposed in existing literatures before. In addition, we have also demonstrated that with different driving and bias conditions, the optical frontend could be re-configured to generate several different advanced multi-level formats, offering flexibility for the system design in dynamic optical networks. Moreover, since the configuration just requires feeding binary electronics, which are widely deployed in traditional systems, it provides an alternative approach to upgrade the existing systems for providing advanced multi-level formats in the physical layer.
The authors wish to thank Dr. A. Chiba of Shizuoka University, Mr. M. Sudo of Sumitomo Osaka Cement and Dr. A. Kanno of National Institute of Information and Communications Technology for their fruitful discussion.
References and links
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